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LLC resonant converter

LLC resonant converter

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lightbulbAbout this topic
An LLC resonant converter is a type of power converter that utilizes a resonant circuit comprising inductors and capacitors to achieve high efficiency and low electromagnetic interference during the conversion of electrical energy. It operates by resonating at a specific frequency, allowing for soft-switching techniques that minimize switching losses.
lightbulbAbout this topic
An LLC resonant converter is a type of power converter that utilizes a resonant circuit comprising inductors and capacitors to achieve high efficiency and low electromagnetic interference during the conversion of electrical energy. It operates by resonating at a specific frequency, allowing for soft-switching techniques that minimize switching losses.
An LLC Resonant Converter with Double Resonant Tanks for Wide-Input-Voltage-Range Applications
by Ezekiel Bokolonga

2025, World Congress on Electrical Engineering and Computer Systems and Science

This paper presents an LLC resonant converter with double resonant tanks configured for a wide-input-voltage range application. The converter comprises four power switches (MOSFETs), a pair of resonant tanks with LLC configuration, and... more
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An LLC Resonant Converter with Double Resonant Tanks for Wide-Input-Voltage-Range Applications
by Ezekiel Bokolonga

2024, Proceedings of the 3rd World Congress on Electrical Engineering and Computer Systems and Science

This paper presents an LLC resonant converter with double resonant tanks configured for a wide-input-voltage range application. The converter comprises four power switches (MOSFETs), a pair of resonant tanks with LLC configuration, and... more
LCC and LLC type resonan conver ers solve the issue depicted by series and parallel configurations, however, both can still portray important drawbacks. Since an optimized design for a resonant converter normally operates close-to/at the resonant frequency, varying the situation when operating at high input vo input voltage also varies the switching frequency, which gives the LCC topology a bad tage. LCC type converter therefore not suitable for this condition since its efficiency degrades due to conduction and switching losses. With this drawback, LLC resonant converter gains the interest and is featured in most recent studies in several three-level and double resonant different types of configurations, ranging from full-bridge, half-bridge, interleaved, anks [2] - [9]. Fi ee a a, i; ee, ee nn i; ik’ c. See Fy ee, ey ee Sy ee, ee ey ee, Sc ee, | en cee on 22k en
Fig. 3: Mode | operating circuits of the proposed converter: (a) Mode 1-I; (b) Mode 1-II; (c) Mode 1-H]; (d) Mode 1-IV.
Fig. 4: Mode 2 operating circuits of the proposed converter: (a) Mode 2-I; (b) Mode 2-II; (c) Mode 2-III; (d) Mode 2-IV. 4 oe Mode 2-II [ti — tz]: This interval starts with magnetizing currents (izm1 and izn2) equal to the resonant currents (iz,1 and iz2) at ti. Diodes D; and D3 on the secondary side both reverse biased, therefore turned off respectively with ZCS. Both magnetizing inductors (Lm; and Enz) involved in resonance with the resonant elements (Lr, Ci, L2, and C2), leaving the secondary side isolated from the primary side. It is the output capacitor C,.; that supplied power to the load. This interval ends with switches S; and Sy turns off at f2, allowing resonant currents to discharge and charge both the output capacitors of switches S> and S23 (Cy.0 and Cy.63) and switches S; and Sa (Coes) and Cocca).
Fig. 5: (a) Equivalent circuit with AC load. (b) Mode 1 AC equivalent circuit. (c) Mode 2 AC equivalent circuit.
In the circuit definitions for (1) and (2), the equivalent resistance Ra- used in Modes 1 and 2 analyses are different, however the value of Q obtained for each mode is the same due to the characteristic impedance of the resonant tank changes with respect to each mode of operation. The turn’s ratio n (N,/N;) of the transformers remains the same in either of the modes.
The performance and operating principles of the proposed converter was verified by a 480 W prototype using the components shown in Table 1. A switching frequency of 110 kHz was used to regulate the output voltage at 35 V, with a fixed duty ratio of 50% on each respective switches, depending on the type of mode used. Combining the input voltages specified in table J for each mode gives the input voltage range of 115 — 230 V for the experiment. This proves the ability of the overall design to operate at an input voltage range of 115 — 460 V, with nominal inputs of 190V and 380V for Modes 1 and 2. Due to limitation of DC input power supply, the prototype was tested only for 115 V (Mode 1) and 230 V (Mode 2) input voltages, giving an output power of 255 W. However, these two inputs provide enough information to validate the design specification. Note that for the overall design, Modes 1 and 2 minimum input voltages are 115 V and 230 V, which is also the voltage levels used in the experiment for each respective mode. Fig. 6 and 7 depicts the key steady state experimental waveforms obtained for the two input voltage levels. Fig. 6 shows the key steady state experimental waveforms of Mode | voltage. By looking at the waveforms in Fig. 6(a), S2 and S3 operate under ZVS condition. Fig. 6(b) shows the waveforms of resonant voltage and current of capacitors (C;1 and C,2) and inductors (L,; and L,2) of t he resonant tanks. The waveforms illustrate the sinusoidal swing of resonant currents iz,1 and iz,2 in opposite directions respectively. This indicates the charging and discharging of resonant capacitors, which depicts symmetrical but due to the resonan he secondary diodes conducted in he distribution of power twice every cycle to the output. The resonating behaviour of these waveforms are expected to be components miss-matched, the amplitudes and phases are not identical. In response, the waveforms of the rectifier diode currents in the secondary side are not perfectly coupled, as shown in Fig. 6(c). However, expected combination. The drain-source voltage of switch S3 and diode current ip; are compared in Fig. 6(c) to show tha ZCS of ip: has been achieved. Fig. 6(d) shows the primary and secondary voltages of ransformers TX, and TX2. From the waveforms, opposite rise and fall of the voltages have been obtained respectively with he secondary side voltages being scaled down by approximately 2 for both primary voltages. With the output voltage of 35 V, the maximum efficiency of the converter for Mode 1 is 96.99%.
Fig. 7 shows the key steady s and Fig. 7(b), Si, S2, S3 and S4 operate under ZVS condition. However, from Fig. 7(b), there is ringing on Vasz and Vas3 due to the output capacitances of S2 and S3 being resonating with the parasitic inductances of the circuit, which also affects Vas1 and Vasa being unbalanced. Fig. 7 inductors (Z,1 and L,2) of both res iz2 in the same direction respectively. This indicates simultaneous charging and discharging of resonant capacitors, which depicts the distribution of power symmetrical, which illustrates ba ate experimental waveforms of Mode 2 voltage. As shown in the waveforms of Fig. 7(a c) shows the waveforms of resonant voltage and current of capacitors (C1 and C2) and pective tanks. The waveforms illustrate the sinusoidal swing of resonant currents iz,1 and twice every cycle to the load. The resonating behaviour of these waveforms is almos ancing mechanism of the converter. In response, the waveforms of the rectifier diode currents in Fig. 7(d) and Fig. 7(e) are likely to be coupled and conducted in expected combination. The drain-source voltage of switches S3 and S4 are compared with diode currents ip; and ips to illustrate the effect of ZCS. Fig. 7(f) shows the primary and secondary voltages of transformers TX; and TX2. From the waveforms, simultaneous rise and fall of the voltages have been obtained respectively with the secondary side voltages being scaled down by approximately 2 for V,,i1 and
(f) Primary and secondary voltages of the transformers.
Table 1: Component list.
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An LLC Resonant Converter with Double Resonant Tanks for Wide-Input-Voltage-Range Applications
by Ezekiel Bokolonga

2024, World Congress on Electrical Engineering and Computer Systems and Science

This paper presents an LLC resonant converter with double resonant tanks configured for a wide-input-voltage range application. The converter comprises four power switches (MOSFETs), a pair of resonant tanks with LLC configuration, and... more
LCC and LLC type resonan conver ers solve the issue depicted by series and parallel configurations, however, both can still portray important drawbacks. Since an optimized design for a resonant converter normally operates close-to/at the resonant frequency, varying the situation when operating at high input vo input voltage also varies the switching frequency, which gives the LCC topology a bad tage. LCC type converter therefore not suitable for this condition since its efficiency degrades due to conduction and switching losses. With this drawback, LLC resonant converter gains the interest and is featured in most recent studies in several three-level and double resonant different types of configurations, ranging from full-bridge, half-bridge, interleaved, anks [2] - [9]. Fi ee a a, i; ee, ee nn i; ik’ c. See Fy ee, ey ee Sy ee, ee ey ee, Sc ee, | en cee on 22k en
Fig. 3: Mode | operating circuits of the proposed converter: (a) Mode 1-I; (b) Mode 1-II; (c) Mode 1-H]; (d) Mode 1-IV.
Fig. 4: Mode 2 operating circuits of the proposed converter: (a) Mode 2-I; (b) Mode 2-II; (c) Mode 2-III; (d) Mode 2-IV. 4 oe Mode 2-II [ti — tz]: This interval starts with magnetizing currents (izm1 and izn2) equal to the resonant currents (iz,1 and iz2) at ti. Diodes D; and D3 on the secondary side both reverse biased, therefore turned off respectively with ZCS. Both magnetizing inductors (Lm; and Enz) involved in resonance with the resonant elements (Lr, Ci, L2, and C2), leaving the secondary side isolated from the primary side. It is the output capacitor C,.; that supplied power to the load. This interval ends with switches S; and Sy turns off at f2, allowing resonant currents to discharge and charge both the output capacitors of switches S> and S23 (Cy.0 and Cy.63) and switches S; and Sa (Coes) and Cocca).
Fig. 5: (a) Equivalent circuit with AC load. (b) Mode 1 AC equivalent circuit. (c) Mode 2 AC equivalent circuit.
In the circuit definitions for (1) and (2), the equivalent resistance Ra- used in Modes 1 and 2 analyses are different, however the value of Q obtained for each mode is the same due to the characteristic impedance of the resonant tank changes with respect to each mode of operation. The turn’s ratio n (N,/N;) of the transformers remains the same in either of the modes.
The performance and operating principles of the proposed converter was verified by a 480 W prototype using the components shown in Table 1. A switching frequency of 110 kHz was used to regulate the output voltage at 35 V, with a fixed duty ratio of 50% on each respective switches, depending on the type of mode used. Combining the input voltages specified in table J for each mode gives the input voltage range of 115 — 230 V for the experiment. This proves the ability of the overall design to operate at an input voltage range of 115 — 460 V, with nominal inputs of 190V and 380V for Modes 1 and 2. Due to limitation of DC input power supply, the prototype was tested only for 115 V (Mode 1) and 230 V (Mode 2) input voltages, giving an output power of 255 W. However, these two inputs provide enough information to validate the design specification. Note that for the overall design, Modes 1 and 2 minimum input voltages are 115 V and 230 V, which is also the voltage levels used in the experiment for each respective mode. Fig. 6 and 7 depicts the key steady state experimental waveforms obtained for the two input voltage levels. Fig. 6 shows the key steady state experimental waveforms of Mode | voltage. By looking at the waveforms in Fig. 6(a), S2 and S3 operate under ZVS condition. Fig. 6(b) shows the waveforms of resonant voltage and current of capacitors (C;1 and C,2) and inductors (L,; and L,2) of t he resonant tanks. The waveforms illustrate the sinusoidal swing of resonant currents iz,1 and iz,2 in opposite directions respectively. This indicates the charging and discharging of resonant capacitors, which depicts symmetrical but due to the resonan he secondary diodes conducted in he distribution of power twice every cycle to the output. The resonating behaviour of these waveforms are expected to be components miss-matched, the amplitudes and phases are not identical. In response, the waveforms of the rectifier diode currents in the secondary side are not perfectly coupled, as shown in Fig. 6(c). However, expected combination. The drain-source voltage of switch S3 and diode current ip; are compared in Fig. 6(c) to show tha ZCS of ip: has been achieved. Fig. 6(d) shows the primary and secondary voltages of ransformers TX, and TX2. From the waveforms, opposite rise and fall of the voltages have been obtained respectively with he secondary side voltages being scaled down by approximately 2 for both primary voltages. With the output voltage of 35 V, the maximum efficiency of the converter for Mode 1 is 96.99%.
Fig. 7 shows the key steady s and Fig. 7(b), Si, S2, S3 and S4 operate under ZVS condition. However, from Fig. 7(b), there is ringing on Vasz and Vas3 due to the output capacitances of S2 and S3 being resonating with the parasitic inductances of the circuit, which also affects Vas1 and Vasa being unbalanced. Fig. 7 inductors (Z,1 and L,2) of both res iz2 in the same direction respectively. This indicates simultaneous charging and discharging of resonant capacitors, which depicts the distribution of power symmetrical, which illustrates ba ate experimental waveforms of Mode 2 voltage. As shown in the waveforms of Fig. 7(a c) shows the waveforms of resonant voltage and current of capacitors (C1 and C2) and pective tanks. The waveforms illustrate the sinusoidal swing of resonant currents iz,1 and twice every cycle to the load. The resonating behaviour of these waveforms is almos ancing mechanism of the converter. In response, the waveforms of the rectifier diode currents in Fig. 7(d) and Fig. 7(e) are likely to be coupled and conducted in expected combination. The drain-source voltage of switches S3 and S4 are compared with diode currents ip; and ips to illustrate the effect of ZCS. Fig. 7(f) shows the primary and secondary voltages of transformers TX; and TX2. From the waveforms, simultaneous rise and fall of the voltages have been obtained respectively with the secondary side voltages being scaled down by approximately 2 for V,,i1 and
(f) Primary and secondary voltages of the transformers.
Table 1: Component list.
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Grid-connected photovoltaic micro-inverter with new hybrid control LLC resonant converter
by huang qisheng

2023, IECON 2016 - 42nd Annual Conference of the IEEE Industrial Electronics Society

 Users may download and print one copy of any publication from the public portal for the purpose of private study or research.  You may not further distribute the material or use it for any profit-making activity or commercial gain ... more
Fig.1 System block diagram of PVMI
Zero voltage switching (ZVS) for the primary side switches and zero current switching (ZCS) for the secondary side diodes can be realized via resonance existing in LLC resonant converter. As shown in Fig. 2, vertical and horizontal axis represents the switching frequency f;, and the phase-shift duty cycle D respectively. Power can be converted efficiently under narrow-range input voltage with control I 1.e. traditional PFM, where the voltage gain is only regulated with f, | 71. Great challenges in magnetic components, system control, and system stability occur when applying control I to PV panels because of the wide-range switching frequency resulted from wide-range input voltage of PV panel and wide-range load variation. Efficient power conversion under wide-range input voltage can be realized through applying control II. As depicted in Fig.2, PFM is adopted in a region where f, is smaller than a specific value, while higher f; for regulation is needed, PS-PWM is used , and so f; range is limited. The efficiency, however, will decrease under PS-PWM on occasions of high input voltage or light load due to the large D which tending to result in the increases of internal circumfluent current loss. Fig.2 Control scheme for full-bridge LLC resonant converter
2. The effect of PS-PWM on limiting switching frequency will be weakened with the increases of Q. It means PFM control performs better than PS-PWM control under heavy loads. Fig.3. fvs. Q and D @ k=3.7 under PFM+PS-PWM mode
Fig.4 Program flowchart of LLC converter with hybrid control The program flowchart of the hybrid control is shown in Fig.4, which is described step-by-step as following. The output voltage Vi; of the LLC resonant converter is compared with the reference voltage in analog to digital (AD) interrupt; thus obtain error Voe70,. The converter will be controlled with PFM if the new calculated switching frequency f; through PI controller is smaller than resonant frequency /;. If not, the PS-PWM+PFM control mode will be activated. Then a modification of phase-shift duty cycle 4D can be calculated according to frequency error which equaling to the differences between f,and f;. Finally, f; and D will be updated by adding AD to a reference phase-shift duty cycle Dpase which had been pre-stored in the register.
Fig.6 M vs. k and Q under PFM+ PS-PWM mode As shown in Fig.5, Jo rms’, io of VS. k and Q in the LLC resonant converter operating PFM mode with f=0.8 are plotted, respectively. Here Jg ms and ig oare normalized as Jo jms" and ig of - It is shown that large k and Q will achieve small currents, then resulting in the low switches loss. And M will be decreased with increasing of & and Q, resulting in wider switching frequency range. Under PFM+PS-PWM mode, the currents behavior is the same like that of PFM mode. In contrast, its M will be increased with increasing of k, as shown in Fig.6. That is to say, large & will not result in wider switching frequency range. Therefore, in the design k and Q of resonant tank, only effect of A and Q on M under PFM control mode is considered.
Fig.7 k, O area meeting the voltage gain and ZVS k and Q, shown as overlapped parts in Fig.7, satisfies both Inequality (2) and (4) . Then finding & and Q to maximize the multiplication of & and Q can achieve optimal design.
f=83.8kHz @ input voltage 39V (under PFM +PS-PWM mode) Using the control scheme and design parameters, simulations of the PVMI are carried. Some key waveforms are shown in Fig.8. vgs, Vas, ip, Vp, V4B, itr represent drive signal of MOSFET, voltage across drain and source, current and voltage of rectifier diode, voltage across A and B in Fig.1, current of resonant inductor, respectively. It is can be seen that, the LLC converter operates with resonant frequency of 65.5 kHz at nominal input voltage 31V and full-load, while operates with PFM+PS-PWM mode, and f;= 83.8 kHz, D=0.2 at input voltage of 39V and full-load. The switching frequency ranges from 47.3 kHz to 111 kHz. The MOSFETs operate with ZVS under full input voltage and loads. It should be noted that ZCS for the diodes, which is in favor of suppressing voltage spike, is also achieved even when f; is higher than f,. The reason is hat the diodes current lead because of phase shift. This ZCS behavior is great different with that under single PFM control .The simulation has good agreement with the design. Fig.8.Key waveforms of LLC converter vs. input voltage
f=65.5kHz@ input voltage 31V (under PFM mode)
4.2 PVMI prototype
fs = 49.1kHz @ input voltage V,y= 25V, output power P,=250W
Jf: =63.6kHz @ input voltage V,,= 31V, output power P,=250W
The MPPT key waveforms are given in Fig.13. Under the test, the Regatron TCP.16.800.400.PV is used as PV simulator. The maximum power point is set to Pyp=250W and Vyp=36V. vpy, ipy , ppv iS output voltage, current and power of the
As shown in Fig.10, efficiency of the full bridge LLC converter is above 96% within MPPT voltage range from 25 to 39V and peak efficiency is high up to 97.7%. Fig.10 Efficiency of the LLC converter vs. V,,, and P,
simulator, and ig,iqis the injected current of the inverter. The test shows that the inverter can search to the maximum power point quickly, and has good MPPT accuracy up to 99.3%. Fig.12 efficiency of the PVMI vs. V,@ Vgrid=220Vac
Fig. 13 MPPT key waveform @ Pyp=250W, Viyp=36V V. Conclusion The PVMI will meet more challenge of achieving high efficiency because of wide-range input voltage, A PVMI with two high-frequency power stages is proposed, and the full bridge LLC resonant converter with hybrid control is used for the first power stage to meet PV wide range input voltage. The LLC resonant converter operates PFM mode when switching frequency is lower than the resonant frequency; otherwise, it
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Multi-phase interleaved LLC-SRC and its digital control scheme
by Hee-Jun Kim

2022, SPEEDAM 2010

This paper proposes a new 3-phase interleaved LLC resonant DC-DC converter and its control scheme. The proposed circuit consists of three conventional interleaved LLC resonant DC-DC converters and each converter operates with π/3 of phase... more
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A ripple reduction method for a two stages battery charger with multi-winding transformer using notch filter
by Haimeng Wu

2022, 2017 IEEE 12th International Conference on Power Electronics and Drive Systems (PEDS)

This paper presents a two-stage battery charger consisting of a bridgeless Totem-pole power factor correction (TP-PFC) circuit and a full bridge converter with a multiwinding transformer. By using this transformer the cell equalizing... more
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ZVS Auxiliary Circuit for a 10 kW Unregulated LLC Full-Bridge Operating at Resonant Frequency for Aircraft Application
by Miroslav Vasic

2022, Energies

In modern aircraft designs, following the More Electrical Aircraft (MEA) philosophy, there is a growing need for new high-power converters. In this context, innovative solutions to provide high efficiency and power density are required.... more
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Hydrogen production system with fuzzy logic-controlled converter
by Salih NACAR

2021, TURKISH JOURNAL OF ELECTRICAL ENGINEERING & COMPUTER SCIENCES

Electrolyte current must be controlled in the water electrolysis systems. For this purpose, the power converter for the cell stack of the electrolyzer used in industrial hydrogen production is realized. A series resonant converter, which... more
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PARTICLE SWARM OPTIMIZATION BASED PID CONTROLLER FOR MPPT
by Petchinathan Govindan

2021, International Journal of Mechanical Engineering and Technology

This work manages an enhanced (NOSLC) which permits signi converter gives high negative output by utilizing the Voltage lift technique. contribution to the converter is given by Photovoltaic module. The PV cells must be worked at their... more
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COMPUTATIONALLY EFFICIENT ROBUST MODEL PREDICTIVE CONTROL OF MAGNETIC SUSPENSION SYSTEMS
by IAEME Publication

2021, IAEME PUBLICATION

Robust model predictive control (MPC) of a general magnetic suspension system (MSS) is analyzed. Tight positioning requirements in the presence of constraints makes MPC a promising control technique for this system. While inherent... more
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COMPUTATIONALLY EFFICIENT ROBUST MODEL PREDICTIVE CONTROL OF MAGNETIC SUSPENSION SYSTEMS
by IAEME Publication

2021, IAEME PUBLICATION

Robust model predictive control (MPC) of a general magnetic suspension system (MSS) is analyzed. Tight positioning requirements in the presence of constraints makes MPC a promising control technique for this system. While inherent... more
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PERFORMANCE ANALYSIS OF LLC RESONANT CONVERTER INCORPORATING PI FUZZY AND NEURAL CONTROLLERS
by IAEME Publication

2021, IAEME PUBLICATION

The existing literature predominantly concentrates on the design of the modified LLC type of resonant converter and the implementation of state space modeling. In this paper, several performance inquiries of PI, Fuzzy and Neural control... more
Fig 1. Circuit diagram of modified LLC Resonant - Boost Converter
Figure 2 Block diagram of LLC Resonant Converter Fed Resistive Load using PI Controlle The Proportional integral (PI) controller is amongst the conventional controllers. It is mostly used to control the speed. The major feature of the PI controller is the capability to keep constant zero steady-state error. The combination of PI is essential to raise the speed of the system response and to reduce the steady-state error. The circuit errors are reduced by trial and error method. The controlled signals are given to RC and Linear transformer [15]. The Pl controllers are a special case; they are usually used for controlling various systems. It has some errors such as proportional and integral errors. Then the output is increased by terms of proportional constant Kp. The integral error also affects the output values [16-17].The output is increased in terms of integral constant Kj. The gain K,and K; are correcting ports to modify the preferred output response. The PI controller has the advantages of increasing system stability. Figure 2 Shows the Block diagram of LLC Resonant Converter Fed Resistive Load using PI Controller.
Figure 3 Block diagram of LLC Resonant Converter Fed Resistive Load using Fuzzy Controllet es, ee FLC works under the principle of FUZZY logics. Fuzzy logic provides conventional and us« friendly frontend to develop the control programs. Fuzzy controllers have the tendency 1 oscillate around the final operating point [18].It is the processes of converting the crisp dat into a linguistic member function. For the given input and output the mapping is formulate: There are seven linguistic member functions. They are NB, N, NS, ZE, PS, P, PB. It is dor by using Takagi-Suge no method [19-20]. Fuzzy logic is a very powerful tool and successfi implementation in every field. Fuzzy controller has several advantages over PI controller. Th fuzzy controller is more beneficial because of its simple design, robust, cheap, and multip! inputs and multiple outputs. Figure 3 Shows the Block diagram of LLC Resonant Convert Fed Resistive Load using Fuzzy Controller.
Figure 4 Block diagram of LLC Resonant Converter Fed Resistive Load using Neural Controller Figure 4 Block diagram of LLC Resonant Converter Fed Resistive Load using Neural Controller output data are important to obtain the desired output values. The data set is adequate and enough to confirm the constant operation. So there is no requirement of erudite after operation. In the Neural Controller the feedback system and pre-filters are multilayered. The Neural controller has different models such as Single input and multiple input models. The rear transmission procedure is used to run the constructed network. This procedure is commonly controlled transmission system for multilayered network. Figure 4 Shows the Block diagram of LLC Resonant Converter Fed Resistive Load using Neural Controller.
Figure 5 Simulated voltage response of LLC-RC under nominal load 100Q and input voltage 40V.
Figure 6 Simulated current response of LLC-RC under nominal load 100Q and input voltage 40V.
Figure 9 Simulated voltage response of LLC-RC under load disturbance (100Q-90Q-100Q) at t=0.55sec. Under Load Disturbance
Figure 8 Simulated current response of LLC-RC under line disturbance (40V-38V-40V) at t=0.55sec. Figure 7 Simulated voltage response of LLC-RC under line disturbance (40V-38V-40V) at t=0.55sec.
Under Line Disturbance
Under Nil Disturbance Figure 11 Simulated voltage response of LLC-RC under nominal load 100Q and input voltage 40V. 4.2. Fuzzy Controller
Figure 12 Simulated current response of LLC-RC under nominal load 100Q and input voltage 40V.
Under Line Disturbance
Under Nil Disturbance figure 17 Simulated voltage response of LLC-RC under nominal load 100Q and input voltage 40V.
Figure 18 Simulated current response of LLC-RC under nominal load 100Q and input voltage 40V.
Figure 22 Simulated current response of LLC-RC under load disturbance (100Q-90Q-100Q) at t=0.55sec.
Table 1 Comparison of various parameters of different controllers under nominal load 100Q.
Figure 23 Comparison graph of different parameters of various controllers under nominal load 100Q.
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POWER FACTOR CORRECTION BY BRIDGELESS BUCK BOOST CONVERTER
by IAEME Publication

2020, IAEME Publication

In current Scenario, efficiency and economic are the major concerns in designing and developing low-power applications. The aim of the paper is to design a Bridge Less Power Factor Corrected (BLPFC) buck boost converter for low power... more
The block diagram of the proposed converter is shown Fig. 1.The proposed converter has two parts. The first part consists of bridgeless rectifier and the second part consists of boost converter with feedback control. Bridgeless buck rectifier consists of two buck converters and these two buck converters are connected in voltage double manner. That means the two buck converters are connected in a parallel input series output manner .The output of this rectifier is given to a boost converter for maximizing the voltage level.
Figure 3 Operation of the Circuit during the Period When Line Voltage is Positive
2.2. Control Strategy of the Novel Three-Phase Bridgeless Boost PFC
The Simulation diagram of the PFC- Buck Boost Converter is as shown in the Fig. 5. Figure 5 Simulation Diagram of PFC- Buck Boost Converter 4. SIMULATION RESULTS
Figure 6 Output of BLPFC- Buck Boost Converter Figure 7 Input Voltage and Current
Table 1 Simulation Parameters
Figure 10 THD Analysis for 90V Input AC Voltage
Dr. B. Vaikundaselvan, Dr. N. Prakash, Dr. S.S. Sivaraju and S. Ashokkumar
The overall hardware diagram for the hardware implementation of power factor corrected by bridge less buck boost converter is as shown in the Figure 12. The high frequency transformer is supplied energy to the converter circuit. The PWM generator is connected the dsPIC Microcontroller. The output is displayed in the LCD display unit. A 12V battery is used in this circuit, one of the single phase inverter output is connected to the Battery. The other inverter output is connected to the DC motor load.
Figure 14 Source Voltage and Current after Converter ON
Dr. B. Vaikundaselvan, Dr. N. Prakash, Dr. S.S. Sivaraju and S. Ashokkumar
Figure 17 Source Current and FFT before Converter Switched ON
The source voltage waveform and corresponding FFT is also shown in Fig. 18. Figure 18 Source Voltage and FFT
Figure 19 Inductor Current and Source Current
Table 2 Output Parameter Comparison
Table 3 THD Measurements of Different Input AC Voltages
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EFFICIENCY IMPROVEMENT USING PHASE SHIFT FULL BRIDGE CONVERTER
by IAEME Publication

2020, IAEME Publication

A server power system is generally composed of three different parts, a power factor correction (PFC) stage, a front-end dc/dc converter stage, and a point of load (POL) stage. The PFC stage is used to achieve unity power factor of the... more
Figure 1 Shows a typical structure of a server power supply.
Figure 2. Circuit diagram of Flyback converter The flyback converter is used in both AC/DC and DC/DC conversion with galvanic isolation between the input and any outputs. More precisely, the flyback converter is a boost converter with the inductor split to form a transformer, so that the voltage ratios are multiplied with an additional advantage of isolation. When driving for example a plasma lamp or a voltage multiplier the rectifying diode of the boost converter is left out and the device is called a flyback transformer.
The forward converter is a DC/DC converter that uses a transformer to increase or decrease the output voltage (depending on the transformer ratio) and provide galvanic isolation for the load. With multiple output windings, it is possible to provide both higher and lower voltage outputs simultaneously. While it looks superficially like a flyback converter, it operates in a fundamentally different way, and is generally more energy efficient.
3. CONVENTIONAL METHOD 3.1. Phase Shift Converter Standby flyback Structur Fly-back converter is the most commonly used SMPS circuit for low output powe applications where the output voltage needs to be isolated from the input main supply. Th output power of fly-back type SMPS circuits may vary from few watts to less than 100 watts The overall circuit topology of this converter is considerably simpler than other SMP, circuits. Input to the circuit is generally unregulated dc voltage obtained by rectifying th utility ac voltage followed by a simple capacitor filter. The circuit can offer single or multipl isolated output voltages and can operate over wide range of input voltage variation. In respec of energy-efficiency, fly-back power supplies are inferior to many other SMPS circuits but it simple topology and low cost makes it popular in low output power range.
4. PROPOSED METHOD A new standby structure where a forward converter is integrated with a dc—dc phase-shift full bridge (PSFB) converter for server power supplies. While a typical standby structure consists of an independent flyback converter, the proposed standby structure is integrated with one leg of the dc-dc PSFB converter. Its main advantages are that the voltage across a standby switch is clamped at the link voltage across the link capacitor and that the standby switch achieves zero-voltage switching over entire load range.
A new integration approach where the reduction of the number of switching components is achieved by utilizing the same switching components in both the dc— dc stage and the standby stage. Specifically, in the proposed standby structure, overlapping each side leg is applied to implement the integration of the dc—dc PSFB converter and the standby two-switch flyback converter, as shown in Fig. The proposed standby structure consists of link voltage
Figure 7. Interfacing the electrical circuit with simulink The Voltage Measurement block acts as an interface between the Sim Power Systems blocks and the Simulink blocks. For the system shown above, you implemented such an interface from the electrical system to the Simulink system. The Voltage Measurement block converts the measured voltages into Simulink signals. Similarly, the Current Measurement block from the Measurements library of powerlib can be used to convert any measured current into a Simulink signal. You can also interface from Simulink blocks to the electrical system. For example, you can use the Controlled Voltage Source block to inject a voltage in an electrical circuit, as shown in the following figure 7.
Figure 8. The above fig shows Simulation circuit for Conventional Phase shift Full Bridge Circuit
Figure 9. The fig shows the simulated I/p V and O/p I,V for Proposed Phase shift Full Bridge Circuit
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IJEET
by IAEME Publication

2020

In the Power management of digital power systems, the advancements in VLSI technology have imposed major challenges such as power density and Hold up time requirement. So in this paper a better design and control Distributed Power System... more
Three designs with different values of inductance ratio (6 = 1, 4,10s) are considered, to evaluate, select and propose a better design for the LLC resonant DC/DC converter. Simulation studies are carried out on these designs to identify a better design with a greater efficiency covering wide range of input voltages. Table 2, represents designed resonant tank parameters of the three designs for the specifications given in Table 1. The resonant tank parameters are calculated using the procedural design steps as given in ce Table 1 Design specifications of LLC resonant DC/DC converter
Table 2 Resonant tank parameters for the three designs 2.1.1. Simulation studies
Table 3 Summary from DC characteristics for the three designs
The mentioned Table 4 gives the details of primary side losses (for a single MOSFET switch), stress on Cr ( V;,) and Stress on secondary side diode (/p) for the three designs at Vin = 400V and V;,, =300V. For the applications in the Front-End DC/DC converter of DPS, the input voltage range is from 300V to 400V with the nominal voltage of 365V. Therefore, the system performance at 365V is also evaluated. The data in Table 3, Table 4 and the system performance at 365V is considered for evaluation. The inferences are cited as under: Table 4 Summary of primary side losses, Stress on C,( V.,) and Stress on secondary side diode ([p) for the three designs at V;,=400V and V;,=300V
Table 5 Details of primary side currents and the corresponding losses for all control Methods Table 5 gives the details of primary side currents and the corresponding conduction and switching loss (Psconduction ANd Pswitching) for a single MOSFET switch at V;,=300V, which is considered as worst operating condition of the converter for the three control methods. This operating condition is considered as the most unfavorable, due to the existence of the highest Root Mean Square (RMS) current on full load. Table 6 Details of secondary side diode currents and the corresponding conduction loss for the control methods
The advancements incorporated in the experimental Set Up are as under. Table 8 List of components used in the experimental set up
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ANALYSIS OF ZVS INTERLEAVED LLC RESONANT CONVERTER FOR CURRENT BALANCING IN DC DISTRIBUTION SYSTEM
by IAEME Publication

2019, IAEME

The power conversion efficiency of an isolated ac-dc converter is a dominant factor in the overall efficiency of dc distribution systems. To improve the power continuity and power conversion efficiency of the dc distribution system, a... more
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FINITE ELEMENT ANALYSIS OF POWER TRANSFORMER
by IAEME Publication

2019, IAEME

Failure in power transformer can be catastrophic to electric systems, since transformers play a vital role in the power sector. Metallic particles in transformer oil lead to Partial discharge which can result in serious conditions. The... more
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FINITE ELEMENT ANALYSIS OF POWER TRANSFORMER
by IAEME Publication

2019, IAEME

Failure in power transformer can be catastrophic to electric systems, since transformers play a vital role in the power sector. Metallic particles in transformer oil lead to Partial discharge which can result in serious conditions. The... more
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Article ID: IJMET_10_02_177 Cite this Article: Chandra Sekhar Garlapati and Ravi Kishore D, Analysis of Zvs Interleaved Llc Resonant Converter for Current Balancing in Dc Distribution System
by IAEME Publication

2019, IAEME

The power conversion efficiency of an isolated ac-dc converter is a dominant factor in the overall efficiency of dc distribution systems. To improve the power continuity and power conversion efficiency of the dc distribution system, a... more
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DESIGN AND ANALYSIS OF FLYBACK MICRO INVERTER FOR INTEGRATION OF FUEL CELLS WITH SINGLE PHASE GRID
by IAEME Publication

2017

Renewable energy is going to play a major role in the future energy scenario. All the existing topologies use number of intermediate stages before conversion of DC input to the AC output to the grid side. For this type of conversion,... more
2. PROPOSED FLY BACK INVERTER SCHEME
The basic operati production of accurate alternating waveform in should be fully disch Seco which must be synch to produce outpu side and to generate output with negative po d produces a sinusoidal wave which can be and ndary side Soi, So2, S03, So4 Must operate p ronized with the grid. S01 inverter output an the grid side i arged before starting of the next switchi roperly to get a and S02 is tu t with positive polarity. S03 and S04 are S tu arity. The out (pu app med to med on to transfer energy circuit smoothens out t filter ng principle that should be followed by the micro inverter for th that the flyback transformer ng cycle. The switches in the ropriate AC waveform transfer energy to gr e id fed to the grid. LO
Figure 2 Illustration of Mode A conduction path When the current goes to zero in the secondary side, there is no voltage reflection across the primary magnetizing inductance of the transformer. As a consequence, the magnetizing inductances begin to resonate with the output capacitor of the primary switch. This mode ends when the primary switch S1 is turned on for the next cycle. The conduction path is illustrated in red in Fig. 5. 3.3. Mode D: t3-t4
Figure 3 Illustration of Mode B conduction path
4, CONVERTER DESIGN
For the simulation of the proposed converter, PSIM ver. 9.1.1 software is used. The proposed converter is simulated in PSIM and the results are obtained from SIMVIEW. Fig. 6 shows the input D.C voltage waveform which is 35.6 V. Fig. 7 shows the waveform of primary switch voltage which is 60 V in steady state. The secondary switch voltage waveform is shown in Fig. 8 and the grid side output voltage waveform which is 230 V r.m.s. is shown in Fig. 9. It can be seen that a pure sine wave output is obtained.
Table 1 Basic specifications of the proposed converter 4.1. Main converter design operate inductan value is For getting a perfect sinusoi the converter in DCM. T transformer primary magne the prim ary side switch as fo it. This will result in high tu tizin in rced g inductan Mm O high peak in primal switch off has to b ff loss in th e switch. en dal wave in the grid side of th he component that ce (Lm ce is high or low, the behaviour of too low, it will result i e converter, it is mandatory to DCM mode of operation is the nding on whether the primary nges. If t and wi ith huge current flowing through the primary inductance ll give an extra stress on inductal nce value is very high, it
Table 2 Specifications of the circuit components The relation between input voltage, turns ratio and peak duty cycle is derived below and expressed in equation (1). Selection of N and Lm is interlinked. These parameters are very much interdependent. When the converter has been designed to operate in the DCM mode of operation it is very important to give sufficient time for complete discharging of the peak magnetizing current of the primary inductance after the primary switch S; has been switched off. In this operation, the worst case is when the converter is operating in maximum duty ratio ‘dy’ and the primary current reaches to the maximum value in the grid cycle. The specifications of all the components used in the converter design are given in Table 2.
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DESIGN OF WIND SOLAR AND PUMPED- STORAGE HYBRID POWER SUPPLY SYSTEM
by IAEME Publication

2017

To rectify the intrinsic defects in wind-solar hybrid system, which is the power generation and electricity load's irrational because of uncertainty of resources. The design of this, can be an agreeable configuration system's capacity... more
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GREENER, SAFER -SUSTAINABLE ENERGY EFFICIENT SHIPPING USING SOLAR POWER OPTIMIZER BY MPPT
by IAEME Publication

2017

In this project work renewable energy solar power free of cost is maximum tracked by SPO & MPPT and gives constant output voltage to meet the energy demand. The shipping industry is facing ever greater challenges, especially regarding... more
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VERSATILE CONTROL USED FOR LLC RESONANT CONVERTER SUITABLE FOR PORTABLE APPLICATIONS
by IAEME Publication

2016

Conventional voltage mode control with one of the Pulse Analog Control Schemes called Pulse-Position Modulation (PPM) offers only limited performance for LLC resonant DC/DC converters undergoing wide variations in operating conditions.... more
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A novel LLC resonant converter with semi bridgeless active rectifier
by Mariusz Bojarski

2016, 2014 IEEE Transportation Electrification Conference and Expo (ITEC)

A novel LLC resonant converter with semi bridgeless active rectifier (S-BAR) is proposed in this paper. The presented secondary side rectifier topology is applicable for EV battery chargers with wide output voltage range. This topology is... more
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HIGH EFFICIENT BRIDGELESS BOOST RECTIFIER FOR LOW VOLTAGE ENERGY HARVESTING APPLICATIONS
by IAEME Publication

2016

A single phase ac-dc bridgeless boost rectifier for low voltage energy harvesting applications is proposed in this paper. The conventional bridge type boost converters for low voltage energy harvesting requires more components hence they... more
A general diagram of an electromagnetic generator is demonstrated in Fig. 1, where k is spring stiffness constant; m is the proof-mass; Dg and Dp represent electrical and parasitic dampers, respectively. Essentially, the energy harvesting system consists of a spring, a proof mass, and an electrical damper. The extrinsic vibrations excite the internal oscillation between the proof mass (magnet) and electrical damper (coils) [2]. Figure 1 General diagram of electromagnetic micro-generator
According to (10), the diode behaves like a controlled power source, the power of which is equals to the power consumption of the equivalent resistor R, (d;).Similar derivations can also be applied to the buck-boost operation cycle [tg, t;), and the same expressions can be derived. So, The switch behaves like an equivalent resistor Re (di) and the diode behaves like a controlled power source, the power of which is equals to the power consumption of the equivalent resistor R, (d;).Thus, the switch network could be modeled by a loss-free resistor and a dependent power source, as illustrated in Fig. 3. Figure 3 Averaged switch model of general two-switch network in Discontinuous Conduction Mode (DCM)
II. Converter Analysis In boost operation mode [0, Tj /2), the instantaneous power delivered to the equivalent resistor in dc equivalent circuit model can be expressed as
voltage. The simplified scheme of the controller and power stage is illustrated in Fig. 5. The converter is designed to operate in DCM [13].
Fig. 14 and Fig.15 demonstrate the boost and buck-boost operations correspondingly. The gate pulses of the switches can be observed in accordance with the input current. Closed-loop voltage control successfully stabilizes the duty cycle at 0.72 at the steady state. As seen from Fig. 16, the output voltage is regulated at 3.3-V dc with approximately 0.2 V (i.e., 12%) voltage ripple. Closed-loop voltage control successfully stabilizes the duty cycle at 0.78 at the steady state. The output voltage for R =300 Q is demonstrated in Fig. 17.
Fig. 13. Diode Voltages and Currents
Table 2 Components and Parameters in the Simulink
Table 3 Theoretical Calculations for Different Loads
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Analysis and Design of an Isolated Single-Stage Three-Phase Full-Bridge with Current Injection Path PFC Rectifier for Aircraft Application
by Sisi Zhao

2016

Nowadays there has been an important increase of electric equipment in modern aircraft which leads to a rising demand of electrical power. This highlights the necessity of efficient, reliable and high energy density power converters which... more
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