Design of electrical machines

A Comparison of Radial and Axial Flux Structures in Electrical Machines

This paper explores the comparative advantages and disadvantages of the simplest form of axial flux geometry, the single sided version, compared to the most common form of radial flux geometry with an inside rotor, when considering... more

We will derive here an estimate of the volume “wasted”, i.e. not used for electromagnetic purposes, in both an axial flux and a radial flux machine rotor. We make a few numeric assumptions for clarity, however the analysis can be redone with different numbers than those assumed here. We begin by assuming machines typically have a flux density in the rare earth magnet of 0.8 T and in the steel of 1.6 T. Under these assumptions then the amount of rotor back iron necessary is Fig. 1: 11 hp (8kW) ic. engine with 8kW single sided axial flux starter alternator attached

Fig. 3. Percentage of volume wasted in a 10 pole radial flux rotor as a function of radius. These values should be compared with the 36% for the typical axial flux machine.

Fig. 2. Percentage of volume wasted in a 50 mm radius radial flux rotor as a function of pole count This is plotted below using some typical values such as ln = 3mm, firstly at a rotor radius of 50 mm, as a function of p, and then at a value of p=10 for a range of rotor radii.

Fig. 4. Airgap area required for a single sided axial flux machine producing 100Nm using an airgap shear stress of 20 kPa, as a function of the ratio of inner to outer radius. Above this are shown the areas required for a range of radial flux machines for the same performance, parametrized by their aspect ratio.

Fig. 5. A cutaway version of the motor on the right in Fig. 4 above. Fig. 4. Pump assemblies, with the original single phase induction machine version on left, PM axial flux version on right

Fasco Australia has recently gone into production of a single sided axial flux, non overlapped winding PM machine with integrated electronics for the swimming pool pump market in Australia. Fig. 4 below shows the 900 W machine as produced, fitted to the wet end, along side the single phase induction machine version it replaces. Fig. 5 shows a cutaway of the assembled axial flux motor. The rotor with magnet segments can be seen, the stator is cut away, but extends from the rotor to the left end shield carrying the bearing, a very small fraction of the volume of the already smaller case. The integrated electronics occupies approximately the same volume as the machine.

Harmonic Impact on Distribution Transformer No-Load Loss

2010, IEEE Transactions on Industrial Electronics, vol. 57, no. 1, pp. 193-200

The losses in European Union distribution transformers are estimated at about 33 TW · h/year, whereas reactive power and harmonic losses add a further 5 TW · h/year. The reduction of distribution transformer no-load loss is particularly... more

Fig. 7. Phase-angle plot of deformed line-to-line and line-to-neutral suppl voltages.

Fig. 11. Wound-core simulated and experimental current waveforms under deformed line-to-neutral voltage excitation.

Fig. 2. Wound-core transformer. (a) One-phase core type. (b) One-phase shell type. (c) Three-phase shell type. the stack core and the wound core. A typical wound core, shown in Fig. 1, is composed of hundreds of long continuous strips of sheet steel. The main advantages include reduction of joints and the use of the grain direction of the steel for the flux path. Wound cores are commonly used for distribution transformers with ratings that range from a few kVA to a few MVA. One- and three-phase wound-core distribution transformers are con- structed by using one or more wound cores assembled about a preformed electrical winding coil, as shown in Fig 2. Il. FE ANALYSIS FOR NO-LOAD LOSS EVALUATION The FE method has been used extensively by many re- searchers for the accurate field analysis of electromagnetic de- vices. A 2-D FE analysis approach used for the analysis of 3-D magnetic components was developed in [19], an FE analysis to validate a new winding design for planar integrated magnetics has been carried out in [20], and a coupled electrothermal

The experimental setup used for evaluating the harmonic impact on wound-core no-load loss is shown in Fig. 3. The block diagram of the experimental setup is shown in Fig. 4. A 23-turn excitation coil was supplied from a single- and three-phase supply via a variable transformer (Regulac, type RQ 25-M, input: 240 V 50/60 Hz, output: 0-275 V, 25 A and Regulac, type RK8-G3M, input: 415 V 50/60 Hz, output: 0-476 V, 8 A) in order to magnetize the wound cores with a distorted supply-voltage waveform. A programmable ac and dc power supply (California Instruments MX30) was used so as to magnetize the wound cores with a sinusoidal waveform.

Fig. 6. Percentage of higher voltage harmonics plot of deformed line-to-line and line-to-neutral supply voltages.

Fig. 8. Wound-core geometry and flux-density vector plot (wound-core width: 190 mm, electrical steel: M-OH 0.27 mm). Fig. 9. Wound-core simulated and experimental current waveforms under sinusoidal voltage excitation.

Fig. 15. Phase-angle plot of experimental and simulated no-load currents under deformed line-to-line supply-voltage excitation.

Fig. 10. Wound-core simulated and experimental current waveforms under deformed line-to-line voltage excitation.

Fig. 17. Phase-angle plot of experimental and simulated no-load currents under deformed line-to-neutral supply-voltage excitation. Fig. 16. Amplitude plot of experimental and simulated no-load currents under deformed line-to-neutral supply-voltage excitation.

Fig. 12. Amplitude plot of experimental and simulated no-load currents under sinusoidal supply-voltage excitation.

Fig. 13. Phase-angle plot of experimental and simulated no-load currents under sinusoidal supply-voltage excitation.

Fig. 14. Amplitude plot of experimental and simulated no-load currents under deformed line-to-line supply-voltage excitation.

By using (12) and the 2-D vector potential, the second left- hand term of (11) is given by Also, the relationship between the differentials of magneti- zation and flux density can be expressed by means of a tensor [16], [17]. Using the Euler scheme to represent the differentials yields

ESTUDO DA CONTRIBUIÇÃO DE UM GERADOR SÍNCRONO VIRTUAL DE GRANDE PORTE PARA A ESTABILIDADE ESTÁTICA E DINÂMICA DE UM SISTEMA DE TRANSMISSÃO

by Patrick Trivilin

In comparison to conventional power plants characterized by Synchronous Generators (GS), Distributed Generation units (DG) usually do not have damping or rotational inertia properties. With the increase of DG sources in the electrical... more

Îndrumar de proiectare Maşini electrice – Proiectarea generatorului sincron

by Spunei Elisabeta and

A computer aided educational tool for design, modeling, and performance analysis of Brushless DC motor in post graduate degree courses

by Yannis L Karnavas

The design of electrical machines is gaining more popularity in the last years especially in electrical engineering courses. However, commonly-used teaching techniques seem to limit students' understanding and prevent the attainment of... more

FIGURE 1 BLDC motor's cross section with detailed geometrical parameters

FIGURE 2 BLDC motor design and performance analysis procedure algorithm developed

Next, for given B,, the produced magnetic flux of each pole (@po1e) can be calculated by applying Equation 8, where Apole 18 the pole's area and can be obtained from Equation 9 Assuming that the magnetic flux is equally spread to both stator and rotor core, the magnetic flux at stator (gy) and rotor (@ry) yoke is derived from Equation 10. Combining Equations 11-15, which define stator slot height (h,,), statot yoke height (iy), rotor yoke height (/,y), flux density at stator yoke (B,y), and flux density at rotor yoke (B,,) respectively, a system with five equations and five unknown variables is created. This system has to be incorporated in the iterative process. At each step of this procedure the variables h,,,, a, and b.s, take new values. The process terminates when the saturation constraints for stator and rotor yoke, which are imposed by material properties and selected by designer, are no longer satisfied. If this happens, the algorithms returns at the starting point of this step and the values of the three aforementioned parameters are properly modified. At this step, except from the constraints which are related to the magnetic circuit of the motor, designer could also incorporate various constraints and requirements, such as the maximum acceptable permanent magnets weight, current density, torque ripple, cogging torque, etc. For the sake of space, these quantities and the relative constraints are not presented here. Reader can refer to Ref. [31] for additional information. Before continuing with the next step, the process involves a last check with regard to the achievement of the required output torque. If this goal has been met, the procedure continues to the forthcoming step (i.e., Step 5). On the contrary, the algorithm returns to Step 3 and then the values of the basic parameters involved in the specific step (i.e., motor's airgap diameter and axial length) are revised. Next, for given B,, the produced magnetic flux of each considered equal to 3.0 for an outer rotor topology. In order to calculate Carter's coefficient slot opening width (b,9) has to be chosen first. In this study, this parameter has been set equal to 2.5mm. Then, the specific coefficient can be derived from Equation 5. In this equation, 7, is the slot pitch and is defined in Equation 6, where kp, is slot opening width to slot top width ratio and is given is Equation 7.

FIGURE 3. General view of the main window of the BLDC designer GUI application along with an exploded view of the menus

the minimum available dc-link voltage. Consequently, the maximum value of back-emf must be chosen with regard to drive system parameters. Then, the number of turns per phase (Npn) is given by Equation 22, where @ is the electrical angular speed. Finally, rms phase current (/,,), conductor's area (Ac), phase resistance (Rp), dc-link current Upc), and dc-link voltage (Vpc) can be obtained from Equations 23-27, respectively. In Equation 25 o¢, donates copper's resistivity and /c,, donates conductor's length.

‘IGURE 4 “Motor Specifications” window—initialization of motor's parameters

2.8 | Step 8—Performance analysis This step involves the calculation of total motor's mass (Mio). The mass of a BLDC motor is comprised of the mass of stator yoke (M,,), stator teeth (M,,), rotor yoke (M.,), stator winding (Mc), and permanent magnets (M,,,). The above quantities can be calculated by applying Equa- tions 28-33, where d,, is stator core density, d,y is rotor core density, dcy is copper winding density and d,,, is permanent magnet density. The losses in the machine windings are pure resistive and can be expressed as shown in Equation 34. The iron losses consist

FIGURE 5 Design parameters and materials properties specification windows of hysteresis and eddy current losses. Assuming a trapezoidal excitation and consequently a trapezoidal flux density variation, the hysteresis losses (Phys) and eddy current losses (Peaay) Per mass unit can be derived from Equations 35 and 36. hysteresis loop. The eddy currents are circulating currents that are induced in the iron core by the magnetic field around the turns of the coil. For a given core, this type of loss is proportional to the frequency, the maximum flux density and the conductivity of the core. Typical values ranges for the hysteresis and eddy current coefficient are 40-55 and 0.04-0.07, respectively for silicon steel laminations. Frictional losses (Pic) have also to be estimated. This type of losses is attributed to the friction between motor's outer diameter (Dou), which can be considered as Dou =2;our and the bearing. They depend on many factors, such as the bearing type, the rotor's outer diameter, rotor's angular speed, and friction coefficient between the rotor's core and the bearings. During motor's starting performance, these losses are negligible and they increase as the speed rises. In general, frictional losses are proportional to the rotor's angular speed. An analytical formula for the frictional losses calculation is given in Equation 37, where k is a constant which takes values between 1.0 and 3.0, M,, is the rotor yoke mass and @ is the hysteresis loop. The eddy currents are circulating currents where k;, is the hysteresis coefficient and k, is the eddy current coefficient for steel, fis the electrical frequency, B is the peak magnetic flux density reached in the core and / is the Steinmetz constant for steel and varies between 1.5 and 2.5 depending on the core material. The hysteresis losses depend on the peak value of magnetic flux density, the frequency of the flux density variation and the area of the

FIGURE 6. General view of proposed tool's post-processing window simulations in order to estimate several electromechanical quantities and motor's behavior at full and no-load operation. Finally, the derived results as well as the motor's configuration are graphically illustrated by the main tool (i.e., using Matlab) for a better perception of the examined problem. For this purpose, various results display capabilities, which can be modified by a dynamic way, are provided to the user. This feature enhances the tool's flexibility and reveals that its main target is not only the accuracy of the calculations during BLDC motor's performance estimation, but also the effective manage- ment and the interpretation of the results in order to maximize the learning benefits. In this way, students are able to validate the design methodology and the results obtained in the lecture assignments and compare these data with the theoretical models, which are provided in the literature sessions. rotor's angular speed. Finally, BLDC motor's efficiency can be written as shown is Equation 38.

FIGURE 7 Graphical results representations from post-processing window (i.e., mesh illustration, flux lines distribution, and flux density distribution of the examined motor

FIGURE 8 Parametric and sensitivity analysis window of the proposed tool

FIGURE 9 Results window (when surface plots is selected) of embrace and magnets thickness variation on: speed, efficiency, cogging torque, magnets weight, input current, output power, output torque, and total motor's weight Furthermore, the proposed here tool enables the conduction of parametric and sensitivity analysis by modifying one or two essential parameters. The above can be done through “Parametric Sweep” window, which is depicted in Figure 8. In the presented case study, user had selected from an available parameters list to vary simultaneously the value of pole arc to pole pitch ratio (for the sake of space also called embrace) and permanent magnet thickness. For this purpose, the initial and stop values of these two variables as well as the variation step have been defined. Then the influence of parameters variation on plenty output variables is investigated. The derived results can be illustrated by two different ways (<e., surface and contour plots). This flexibility permits each student to choose the display capability, which is more effective, allows him to validate the theoretical background and extract more easily useful conclusions. Figures 9 and 10 depict the corresponding results of such a parametric analysis conducted for our case by using surface and contour plots respectively. Finally, by selecting an embrace- magnets thickness combination several plots depicting various electromechanical quantities can be created, as shown in Figure 11. User can select the quantities that will As already mentioned, the described here software program has been developed in Electrical Machines Laboratory of DECE at Democritus University of Thrace. It was intended to provide students an interactive and quite useful tool in order to study and validate BLDC design and performance analysis procedure aiming to stimulate their interest in broaden electrical machines area. For this purpose, this tool was integrated in learning process of the “Advance topics in Electrical Machines Design’ course at post-graduate level

FIGURE 10 Results window (when contour plots is selected) of embrace and magnets thickness variation on: speed, efficiency, cogging torque, magnets weight, input current, output power, output torque, and total motor's weight (MSc and PhD degree). During the spring 13-weeks semester of the 2014-2015, 2015-2016, and 2016-2017 academic year, 36 students attended the specific course. According to the syllabus, the course consists of a 4-hr session in a weekly basis. As it can be seen in Figure 12, where the course schedule is presented, the specific course is organized in three different parts (Part 1-3). The first 4 weeks (Part 1) are devoted to provide students with the necessary theoretical background that governs this topic. Specifically, in these theoretical lectures, a thorough discussion and analysis is conducted pertaining to (a) the fundamental principles of BLDC machines operation; (b) the equivalent electrical and magnetic circuit; (c) the aforementioned design methodology; and (d) the involved equations. At the end of Part 1 (fourth week) a test is administrated to the students. The main purpose of this «pre-tool» test is to assess the students' fundamental knowledge of BLDC machines and ascertain if the theoretical background was well received by them. classroom after the homework's deadline according to the schedule (Figure 12). Except from these weekly projects, one final project (FP) is assigned to each post-graduate student. This project includes the design of a BLDC motor, which is supposed to be suitable candidate for various industrial and electromotive applications. For example, in the last academic year (i.e., 2016-2017) students had to develop a direct-drive traction system for an electric scooter designing a BLDC outer rotor motor with surface-mounted magnets. Taking advantage of the tool's capabilities (e.g., parametric and sensitivity analysis, provided graphical feedback, etc.), students had to ensure that the final motor configuration will satisfy a large amount of constraints in agreement with the specific application requirements. The target of this procedure is the students to conclude to a motor assembly close to the optimal one through the appropriate post- processing analysis and by applying engineering judgment. The final deliverables involve a technical report in which the analysis of the derived motor topology is conducted along with a presentation. All students present their work in the last two sessions of the course (Part 3). Furthermore, a «post-tool» test is administrated to them. This exam involves the same questions with the «pre-tool» test, since the correct answers of

FIGURE 12 “Advance topics in Electrical Machines Design” course's semester schedule and organization

FIGURE 11 Plots window which depicts several electromechanical quantities for a given embrace-permanent magnets thickness combination. From left to right and top to bottom: airgap flux density versus rotor angle, efficiency versus torque angle, cogging torque versus rotor angle, induced phase voltage versus rotor angle, output power versus speed, and efficiency versus speed

FIGURE 13 Overall (3-semester) students’ grade distribution on post-graduate course “Advanced Topics in Electrical Machines Design”

FIGURE 14 Overall performance results of “pre-” and “‘post-” tool test grade distribution for the 36 post-graduate students who attended the specific course over the last three academic years. According to DECE's grading scale system it is supposed that a student fails a course when his grade is less than or equal to 49% of the maximum grade that the student could receive. Thus, in the grading scale of 0-100 points a “Fail” corresponds to a total score less than or equal to 49 points, a “Pass” grade varies from 50 to 64 points, while “Good,” “Very Good,” and “Excellent” performance corre- sponds to 65-74, 75-84, and 85-100 points, respectively. Based on the performance results, only one of the 36 students failed to receive an accessible score as he did not meet the requirements of the final project. On the other hand, 9 students attained “Pass” and “Very Good” as final grade, 13 students attained “Good,” while the performance of four students was considered to be “Excellent,” which corresponds to 11% of To quantitatively assess tool's impact and its acceptance on the above learning process and clarify students’ satisfac- tion level a survey was undertaken at the last session of the course for three consecutive academic years. Students were asked to fill a questionnaire, in which they had to grade the 13 questions form according with the five-level Likert scale, where “1” means strongly disagree, “2” means disagree, “3” means neither agree or disagree, “4” means agree, and “5S” means strongly agree. Thus, they were able to express their opinion about tool's usefulness in consolidating their knowledge of this field. Students' feedback is essential, since

FIGURE 15 Cumulative results of students’ survey great interest by students. According to the results (Questions 1-3), students at first place found that the educational tool is very user friendly, since they mentioned that its interface has been well-designed allowing them to insert easily the required input data and change the geometrical and operational parameters of the machine. The particularly high average attained in Question 3 highlights the effectiveness of the developed graphical interface. Furthermore, it points out that instructors’ great effort so that to enhance tool's flexibility and incorporate various results display capabilities has been positively appreciated by students. At this point, it should also be mentioned that the tool's enhanced display capabilities, the inclusion of parametric analyses ability, the inclusion of a brief help at each menu of the tool and the capability of plots creation which depict simultaneously several electromechan- ical quantities, were implemented in order to help post- cal quantities, were implemented in order to help post- As it can be seen, the average grade of each question is equal or higher than 4.00 in the 1.00-5.00 scale. This feedback indicates quite positive learning experience and

und high efficiency. He has gained a full time award sc- tion machines with characteristics of high power density

TABLE 1 Students’ survey questions and the corresponding results it gives instructors the proper guidance about which characteristics of teaching approach and especially in our case which tool's specifications should be modified or not. Therefore, the conduction of several crucial improvements for forthcoming sessions is enabled. Moreover, the interaction between students and educators ensures the effectiveness of the course, the maintenance of the quality of university education and permits them to reflect upon their experiences among others. The 13 imposed questions and the correspond- ing students’ responses obtained through evaluation sheets are summarized in Table 1. Additionally, the same Table shows the average of each question. The cumulative results of students’ survey are also depicted in Figure 15. characteristics of teaching approach and especially in our

A Practical BLDC Motor Design Procedure for Diver Propulsion Vehicle Applications

by Yannis L Karnavas

In this paper a practical approach for the design and construction of a brushless DC motor (BLDCM) for a diver propulsion vehicle (DPV) is presented. After evaluating the corresponding vehicle's characteristics, a three step process is... more

Fig. 1. A typical commercial DPV in the 300W motor power range. A DPV construction is significantly smaller than their larger counterparts i.e. AUV and much more cost-effective since the price of the most expensive ones can reach 13,000€. A DPV’s external view can be seen in Fig. 1. It consists of the hull and the propeller along with its protector externally, while the batteries, the motor and some control circuits are

Fig. 3. Final propeller visualization result a) OpenProp view, b) Solidworks 3D model using OpenProp results.

Fig. 2. Propeller efficiency results obtained from DPV parametric analysis.

Fig. 4. B-H experiment: a) theoretical circuit, b) laboratory setup, c) B-H loop from processed samples and d) material B-H curve used for the design. voltage period is 20ms, the minimum sampling frequency should be f, = 10 kHz (7, = 0.1 ms for 200 samples/period) and the maximum may be f, = 50 kHz (T, = 20 ps for 1000 samples/period. In this study, the maximum sampling frequency was chosen for more accurate results. It should be noted here that for proper results evaluation, the Ves. Should not be sampled in RMS values and also, the dc component must be eliminated. This can be easily implemented in a post- processing phase by subtracting from each measurement the average value of the total measurements. Further, core losses can be calculated using the formula:

Fig. 6. Flux density as a function magnet height for g=1 mm and N,,=16.

Fig. 7. Parametric investigation a) Airgap flux density as a function of magnet embrace, b) embrace as a function of magnet width.

Fig. 8. FEA results from the designed BLDC motor, a) airgap flux density, b) cogging torque variation.

Fig. 9. Characteristic curves (efficiency, speed, current and output power as a function of torque) of the designed BLDC motor.

ee Ae Fig. 11. Constructed BLDC motor prototype, a) completed stator with three phase windings, b) rotor with magnets in place along with 3D printed spacers (green), c) front view of assembled motor including shaft, d) rear view of the assembled motor mounted in a painted aluminum base. Fig. 10. Designed BLDC motor a) overall geometry and b) magnetic flux density distribution at rated operation.

NOMINAL DATA AND PARAMETERS THE DESIGNED BLDC MOTOR

flux density can be calculated by Eq. (9) where B, is the flux density produced by the magnets, ju, is the relative magnetic permeability and k;,,; is the leakage flux coefficient. Fig. 5. Associated BLDC geometry a) slot and teeth, b) overall radii.

Transformers Made of Composite Magnetic Cores: An Innovative Design Approach

2009, Journal of Recent Patents on Electrical Engineering, vol. 2, no. 1, pp. 1-12,

Nowadays, transformers are made of conventional magnetic cores which are constructed of a single grainoriented or amorphous, magnetic steel. Even though, the transformer is the most efficient of electrical machines, with efficiencies... more

Fig. (1). Treanor’s wound core, extracted from patent US2960756, 1960.

In 1980 Lin, Ellis, and Burkhardt disclosed a composite wound core having two parallel magnetic circuits of unequal mean lengths as shown in Fig. (2) [42]. The short magnetic circuit, i.e. inner part of the core, is constructed of a high permeability grain-oriented steel, having low loss but high cost unit, and the long magnetic circuit, i.e. outer part of the core, is constructed of a conventional grain-oriented steel, having higher loss but low cost unit [42]. Fig. (2). Composite wound core, extracted from patent US4205288 1980.

Fig. (5). Specific core loss curves of conventional, M4 0.27 mm, and high permeability, M-OH 0.27 mm, grain-oriented steels. Fig. (4). Normal magnetization curves of conventional, M4 0.27 mm, and high permeability, M-OH 0.27 mm, grain-oriented steels.

Fig. (3). Peak flux density distribution across the limb of a strip wound core.

High permeability grain-oriented steels present improved magnetization and specific core loss characteristics in comparison with the conventional grain-oriented steel as illustrated in Fig. (4 & 5). On the other hand the cost unit of high permeability steel is higher than that of conventional grain-oriented steel. Thus, by using conventional grain- oriented steel for the inner and outer part of the wound core and high permeability grain- oriented steel for the rest part of the core, the transformer manufacturer may achieve an optimum tradeoff between first cost and PV of future no load loss.

Fig. (6). Representation of the multiple grade lamination wound core geometry and variables.

Fig. (9). Current probe and differential voltage probe.

Figure 10 illustrates a detail of the wound core, the excitation coil, and the search coils. The wire used for the search coils is a solderable enameled copper wire 0.1 mm in diameter. A total of 30 search coils were inserted in the wound core for the experimental evaluation of its peak flux density distribution along the limb, yoke and comer. The overall loss of the core did not change noticeably after the coils were wound in position. Therefore, it was assumed that the flux distribution did not change much due to the search coils being introduced. Fig. (10). Detail of the wound core, the excitation coil and the search coils.

Fig. (8). Experimental setup. The experimental setup used for the local flux density and no load loss measurements of wound cores is illustrated in Fig. (8). It consists of a PC and a National Instruments (NI) 6143 data acquisition (DAQ) card (8 voltage differential inputs, 16 bit, 250 ksamples / s). The excitation coil terminals are connected to the input of the DAQ card via an active differential voltage probe (3 dB frequency of 18 MHz) and a current probe based on the Hall Effect (1 dB frequency of 150 kHz) is used for capturing the no load current. Both probes are depicted in Fig. (9). For the magnetization of the wound cores, a 23 turn excitation coil is supplied by a one- phase 230 V, 50 Hz source through a variable transformer. The no load loss was evaluated by the acquired voltage and current data with appropriate virtual instruments (VI) created with the use of LabVIEW.

Fig. (12). Contours of the magnetic vector potential of a conven- tional wound core.

Fig. (14). Peak flux density nodal plot of a multiple grade lamination wound core(B =1.5T, x, =0 mm, x, =10 mm). Fig. (13). Peak flux density nodal plot of a conventional wound core(B =1.5T).

Fig. (11). Areas comprising the 2D model of a conventional wound core.

Fig. (19). Peak flux density distribution along line AB for B =1.5 T and x, =5 mm, x, =10mm. Fig. (18). Peak flux density distribution along line AB for B =1.4 T and x, =5mm, x, =10mm.

Fig. (15). Peak flux density nodal plot of a multiple grade lamination wound core(B =1.5T, x, =5 mm, x, =10 mm).

Fig. (17). Peak flux density distribution along line AB for B =1.5 T and x, =Omm, x, =10mm.

Fig. (16). Peak flux density distribution along line AB for B =1.4 T and x, =Omm, x, =10mm. Table 1 presents a comparison between the calculated no load loss with the FEM models of Section 4 and the respective experimental values for a number of x,, x, configurations. The considered wound core parameters are X, =24.3 mm, x, =57 mm, x, =183 mm, x, =190 mm, and the mean flux density is B =1.7 T. As can be seen from Table 1, the computed results are in good agreement with the measured ones and the error is less than 5% in all cases. Also the difference between the computed no load loss with the 2D and 3D FEM models is insignificant and is less than 1.1% in all cases.

Fig. (21). Detail of the peak flux density vector plot of the multiple grade lamination wound core with optimum design variables x,,

Fig. (20). Variation of the objective function with respect to SAR iterations. During the minimization of (18) a number of local minimums with values very close to the global minimum were identified. Due to this fact the number of successes of the GA was lower than that of the SA and SAR algorithm, as the GA was trapped most of the times in local minima. The optimum distribution of the design parameters as calculated by the SA and SAR algorithms correspond to an objective function value of f «) = 677.5 $, ie. a 3.2% reduction of the sum of magnetic steel cost and PV of future no load loss as it was pre-evaluated by the industrial optimization scheme [48]. Furthermore, the SAR algorithm exhibits a 20% reduction in objective function calls comparing to the classic SA algorithm. Figure 20 depicts the variation of the objective function with respect to the SAR iterations and Fig. (21) illustrates a detail of the computed flux density distribution with the 3D FEM model of the multiple grade lamination wound core with the optimum variables.

Table 1. Comparison Between Measured and Computed no Load Loss (x, = 24.3 mm) optimization with a heuristic methodology which is already used by the transformer industry [48]. The parameters of the wound cores under consideration are x, =51.3 mm, x, =57

Design and Analysis of Permanent Magnet Synchronous Motor used in Industrial Robot

by ဒေါက်တာအောင်ဇေယျ (Professor Aung Ze Ya)

2011, Proceedings of the Fourth National Conference on Science and Engineering (Northern Myanmar)

This paper presents design and analysis of Permanent Magnet Synchronous Motor (PMSM) used in industrial robot. Robotics is the applied science of motion control for multi-axis manipulators and is a large subset of the field of... more

Design of Slotless Permanent Magnet Induction Synchronous Motor

by Santhosh S

2014, proceeding of NCIEEE'14

The slotless Permanent magnet Ind-syn motor (SPMISM) have been developed in the recent years and gained popularity in various industrial applications. It is quite possible that this SPMISM will become predominant in the near future... more

Thus in order to generate smooth torque at various operating conditions, the current waveform should be sinusoidal and in phase with the BEMF. In slotted motors stator teeth begin to saturate at operating conditions near rated value. This results in both a decreased efficiency due to core losses and a non linear torque per current characteristics. By keeping iron out of the windings, no saturation is present in slotless machine. This results in decreased iron losses, increased efficiency. Fig.2. Cross section of slotless pmism

Fig.3. Schematic arrangement of rotor The rotor has a robust rotor construction and low effective air gap. Operations at high speed are also possible. The SPMISM produces _ sufficient accelerating torque due to the cage winding[8],[9],[11]. Use of PM in the rotor of the SPMISM makes it unnecessary to supply magnetizing current through the stator for constant air gap flux. The SPMISM will operate at a higher power factor because of the absence of magnetizing current. The AC drives like synchronous motors requires DC excitation, which is often supplied through brushes and slip rings. This implies rotor losses and regular slip ring maintenance, which would result in the down time. The SPMISM is free of this drawback because the field coil, DC power supply and slip ring not necessary for the SPMISM.

A SPMISM has slotless stator and cage rotor with PM (NdFeB-neodymium-iron - boron rare earth magnet) Buried in the rotor core[6]. The stator core has no slots. So that the windings are fixed or wounded on stator core using adhesive method or printed circuit board. Fig. 1. Cross section of slotted pmism

The SPMISM starts with dominant induction motor characteristics.[13] When the rotor speed reaches synchronous speed, the field set up by the permanent magnets gets locked with stator magnetic field, making the rotor to run at synchronous speed. As and when the rotor reaches the........... weed synchronous speed, the cage torque is reduced to zero and magnet torque becomes the only source of motor torque .Thus we require smooth self starting and also runs at synchronous speed. The purpose of the slotless stator is to reduce the core losses. volume ratios, linear current versus torque, no commutator or slip rings due to slotless stator and permanent magnet cage rotor design The use of Slotless stator, Permanent rotor construction of electrical machines brings following features:

Fig.5. Analytical method 2.2 kw.4 pole,50Hz,3 phase SPMISM:

The proposed system is very compact and which makes the installation an easy one. The system will operate at optimum efficiency at different loads, which is not at all possible in conventional system. The energy efficiency of the system is high and 25% of energy can be saved. In agricultural sector the system efficiency is ignored by due to ‘free power’ available to this sector at present, and the selection is made mainly based on the initial investment. Now state electricity board (SEB) is looking for electrical tariffs at par with other utilities. With a scope for considerable energy a cost savings available in the proposed technology, the consumer will opt for new and replacement requirements. Further the capital expenditure incurred will be paid back in an attractive periods. Fig. 6.[15] Photos of the slotless PM motor prototype

TABLE 1. COMPARISON OF INDUCTION MOTOR PRESENT AND PROPOSED The performance comparison of three phase SPMISM with conventional induction motor is given in the following table. all the three motors are specification (2kw,4 pole ,50hz,SPMISM). Table 1 clearly emphasizes that the losses produced by my motor SPMISM is comparatively lower than all other motors. The efficiency by the SPMISM is 96.76%. Hence in all accords my motor gains its superiority. Power factor gained from SPMISM is nearer to unity.

Design Methodology of a Switched Reluctance Machine for Off-road Vehicle Applications

by Wasi Uddin

This paper presents the design of a Switched Reluctance Machine (SRM) for off-road hybrid electric vehicle applications. SRMs are ideally suited for this application for their low cost, fault tolerance and reliable operations. A... more

An Investigation Study Considering the Effect of Magnet Type, Slot Type and Pole-Arc to Pole-Pitch Ratio Variation on PM Brushless DC Motor Design

by Yannis L Karnavas

—This paper deals with a sensitivity study of an outer rotor permanent magnet (PM) brushless DC (BLDC) motor design, concerning the effect of several parameters and factors on its corresponding performance. For the purposes of this study,... more

Thermal Investigation of Permanent-Magnet Synchronous Motor for Aerospace Applications

2014, IEEE Transactions on Industrial Electronics, vol. 61, no. 8, pp. 4404-4411

This paper concerns the thermal investigation of a surface-mounted permanent-magnet synchronous motor designed for high-temperature aerospace actuation applications. A finite-element package was developed, enabling accurate 3-D transient... more

Modelling and multiobjective optimization analysis of a permanent magnet synchronous motor design

by Yannis L Karnavas and

An alternative to traditional low-speed/high-torque drive systems, which are currently used in industry, could be the use of a permanent magnet synchronous motor directly coupled to the load and running at low speed, instead of the... more

FIGURE 1 Geometrical representations for the permanent magnet synchronous motor designs under study. A, General cross section (inner rotor). B, Detailed slot geometry (1 pole pair) with inner rotor configuration. C, Detailed slot geometry (1 pole pair) outer rotor configuration The paper aims to provide a rigorous insight in the PMSM design modelling and optimization problem through finite element method (FEM) as well as to investigate the effec- tiveness of specific optimization algorithms regarding their ability to find solutions of PMSM designs (as competitive alternatives) to replace conventional industrial gearbox-low speed drive systems equipped with induction motors. The work performs also a systematic parametric investigation aiming to determine (1) the effects of certain variables vari- ation and (2) the influence of the core material used, to the reached optimized PMSMs. The organization of the work is as follows: The relative geometrical topologies and the nec- essary mathematical background of the overall motor design procedure are presented at Section 2, while the adopted procedure are presented at Section 2, while the adopted In radial flux PMSM, the current flows axially, while the magnetic flux flows radially inside them. Figure | depicts a generic cross section as well as detailed geometrical represen- tations of the configurations under study.*

where py, the relative permeability of the magnets, K jeax express the flux lines flowing through the air-gap (in percent)—called leakage factor—and B, is the magnet’s remanence flux density. By performing magnetostatic FEM simulations, it is easy to obtain the leakage factor. In this way, a simple although accurate linear relationship of k eax regarding the PMSM poles p can be used!”

It should be stated here that, for both configurations, the widths of the slots shown in Equations 4 to 5 and Equations 9 to 10, are considered as straight lines (and not arcs as they compared to the pitch of the slot rt, is large enough.

FIGURE 2 Typical permanent magnet motors drive setup (through the optimization of the rotor geometry) can be jus- tified, by observing Equation 16. Because of the frequency square proportionality of the eddy current losses, the latter dominate in high speeds only. However, the hysteresis losses are prevail here, because the frequency used in our cases is higher than its fundamental component.

FIGURE3 Typical design optimization results obtained. Geometry and flux density distribution for inner rotor topology and CF; objective function: A, Fmincon. B, genetic algorithm

FIGURE 4 Typical design optimization results obtained. Geometry and flux density distribution for outer rotor topology and CF; objective function: A, Fmincon. B, genetic algorithm

IGURE5 B-H curves of the 12 additional (and commercially available) materials used here for permanent magnet synchronous motor stator and rotor cores Abbreviations: FMC, Fmincon; GA, genetic algorithm; PS, pattern search. weight are shown for inner and outer rotor geometries, respec- tively, (for the 3 cost functions and through the 3 algorithms used). It is seen that the quantities ranges (min-max) found are 22 to 122 kg, 71.4% to 89.05%, and 1.9 to 5.3 kg. These tables reveal that if, for example, the machine weight is the pri- mary objective (cost function CF), the outer rotor topology is the choice, with the GA “showing the way” by presenting an efficient (approximately 80%) and very light (approxi- mately 22 kg) motor, while keeping the magnet weight low enough (approximately 4 kg) as seen in Table 4. The second place goes to FMC method applied to the same geometry, while for the PS method, it can be said that fails to present satisfactory results of the same degree. In the case where the magnets weight is the primary objective (cost function CF;3), the same conclusions are true. Indeed, GA presents a solution with quite satisfactory efficiency (approximately 76.4%), low magnet weight (2.4kg), while keeping the total machine weight low (approximately 27.5 kg), as also shown in Table 4. Fmincon provides the second “alternative” again, in Table 4. Fmincon provides the second “alternative” again,

FIGURE 6 Effect of ferromagnetic materials (for stator and rotor cores) on optimized permanent magnet synchronous motor’s (PMSM’s) weight for int and outer rotor topologies (optimization method Fmincon, cost function CF;)

FIGURE7 Effect of variation of pole-arc per pole-pitch ratio # (x-axis to all subgraphs) and magnet length /,, (y-axis to all subgraphs) on 9 quantities c optimized permanent magnet synchronous motor with Fmincon method and cost function CF}. A, Inner rotor topology (see Figure 3A). B, Outer rotor topology (see Figure 4A). PM, permanent magnet The results obtained previously were calculated by incorpo- rating M19_24G as the stator and rotor core material and NdFe35 type for the PMs. M19_24G is a silicon nonori- ented (its magnetic properties are practically the same in any direction of magnetism in the plane of the material) electri- cal steel. It is commonly used in large generator, motors, and transformers, as it has superior permeability at high induc- Finally, Table 6 summarizes the thermal calculations results involving the slot current density for each one of the geometries found. As stated in Section 2.4, only 2 (outer rotor)

FIGURE 8 __ Effect of variation of pole-arc per pole-pitch ratio f (x-axis to all subgraphs) and magnet length /,, (y-axis to all subgraphs) on 9 quantities ¢ optimized permanent magnet synchronous motor with genetic algorithm method and cost function CF3. A, Inner rotor topology (see Figure 3B). B, Oute rotor topology (see Figure 4B). PM, permanent magnet Based on the above, the analysis took place with the appli- cation of 22 different materials. Ten of them were available through the commercial softwares database used, while the rest 12 were processed and incorporated by the authors. For the updating of the softwares database, the authors made an In every case, the same material was tested for both cores. We consider the geometries of the third cost function (CF3) optimized with FMC method (Figures 3A and 4A) since we choose that the magnet weight is of primary concern,

FIGURE9 Effect of variation of pole-arc per pole-pitch ratio # (x-axis to all sub-graphs) and magnet length 1, (y-axis to all sub-graphs) on 9 quantities of optimized permanent magnet synchronous motor with pattern search method and cost function CF. A, Inner rotor topology. B, Outer rotor topology. PM, permanent magnet the predefined efficiency. This is commented as an expected feature since the energy density of the magnets used is high enough. Therefore, Figure 6 depicts only the total machine weight recordings, where the material performance has been sorted out from lower to higher values. It can be seen that there are 9 materials that score lower machine weight val- ues than that of the M19_24G initially used. Based on that, from a manufacturer’s point of view, the material commer- cial cost is another crucial affecting factor in order for the machine designer to make an appropriate choice for a certain application. effort to retrieve real data from commercially available mate- rials in the free market™ and secondly proceed with data entry of the relative B-H curves of these materials as well as their properties, ie, bulk conductivity, mass density, using the software capabilities. For reference reasons, the B-H curves of the new (added) materials (along with their com- mercial names) are shown in Figure 5. By using one of the materials each time, simulations were performed and the val- ues of (1) the efficiency, (2) the output torque, and (3) the total weight of the PMSM were recorded. It was noticed that all of the materials achieved the required torque as well as all of the materials achieved the required torque as well as

Abbreviations: FMC, Fmincon; GA, genetic algorithm; PMSM, permanent magnet synchronous motor; PS, pattern search. TABLE1_ Results of PMSM design variables through FMC, GA, and PS optimization methods (inner rotor) TABLE2 Results of PMSM design variables through FMC, GA, and PS optimization methods (outer rotor) A and A@ are matrices, x, b, b®, 1’, and uw? are vec- tors; C(x) and C(x) are functions, which return vectors; and F(x) is a function that returns a scalar. Finally, F(x), C(x), and C*/(x) can be nonlinear functions. Three optimiza- tion methods were chosen and applied here for solving our problem expressed in the form of Equation 28. These are (1) Fmincon (FMC), (2) genetic algorithm (GA), and (3) pat- tern search (PS). These techniques have been proved robust and efficient enough to most engineering problems in the past. Moreover, since they are well known, their principles of operation have been extensively presented in the litera- ture, and therefore, no relative information is given here. For further implementation details, the reader can refer to the literature.2°*? Among several parameters a GA uses, the most important ones refer to the population size (Ps), the number of generations (G,,), the crossover rate (C;,), and the mutation rate (M,). These parameters are given here the following values: P; = 80, G, = 100, C,=0.75, and M,=0.05. Eighteen sets of d esign results were derived (3 methods x 3 objective functions x 2 rotor topologies). For the inner rotot topology, Table 1 design variables t! shows the optimization results of the 12 hrough FMC, GA, and PS and for each one of the cost functions applied (CF)-CF3). Table 2 shows the same results for t he outer rotor topology. For demonstration purposes, some O f the optimized designs obtained (objective function CF3) were visualized and are depicted in Figures 3 and 4, with respe ct to (w.r.t.) their geometry and their flux density distribution. From these tables and figures, it is ini- tially clear that all algorithms succeeded to satisfy all of the existing constraints and to converge to a-near-optimum desigr solution. As an example, the motor’s outer diameter is kept within satisfactory limits (24-42 cm, with a 20- to 50-cm con- straint). while the machine lenoth lies mainlv in the 10- tc

Abbreviations: FMC, Fmincon; GA, genetic algorithm; PMSM, permanent magnet synchronous motor; PS, pattern search.

TABLE3 Objective quantities results through FMC, GA, and PS optimization methods (inner rotor) Abbreviations: FMC, Fmincon; GA, genetic algorithm; PS, pattern search.

TABLE4 Objective quantities results through FMC, GA, and PS optimization methods (outer rotor) Abbreviations: FMC, Fmincon; GA, genetic algorithm; PS, pattern search.

TABLE5 Computational cost (seconds) of optimization methods applied Abbreviations: FMC, Fmincon; GA, genetic algorithm; PS, pattern search.

TABLE 6 Slot current density (A/cm?) results of the 18 designs

TABLE7 Optimum combinations of embrace and magnet length (first 2 rows) for minimum PM weight and PMSM current and relative performance quantities Abbreviations: FMC, Fmincon; GA, genetic algorithm; PM, permanent magnet; PMSM, permanent magnet synchronous motor; PS, pattern search. 4.3 | Effect of pole-arc per pole-pitch ratio and magnet length been preoptimized already, the desired rated torque and output power are almost always satisfied for any of # and /,, combina- tion. Also, for easy reference purposes, a “dot” sign following by its corresponding value has been assigned to all subgraphs of Figures 7 to 9 . The best combination chosen actually exhibits the minimum motor line current (for maximum eco- nomic operation), while at the same time, the corresponding magnet weight can be considered as the minimum necessary quantity which ensures the desired nominal specifications during this economic operation. From a “decision-making” point of view, it is concluded that the inner rotor configuration preoptimized by GA method (Table 1) combined with best f and /m values of Table 7 presents the most attractive solution among the others. It exhibits low torque angle, low cogging torque, low current density, low magnet weight, low enough machine weight, low iron and copper losses, and the highest efficiency. Finally, it should be noted that the investigation, which took place in this subsection is judged as a necessary stage (refinement) in the modelling and optimization proce- dure proposed here, since the pole-arc to pole-pitch parameter has not been taken into account in the analytical calculation stage described in Section 2 and to which all methods adopted were based on.

TABLE A3__ Values of the constants involved in the design procedure

TABLE A2___ Design search variables and their universe of discourse

Abbreviation: PMSM, permanent magnet synchronous motor. TABLE A4_ Constraints of the PMSM design problem

Optimizarea proiectării unui generator sincron ce echipează o microhidrocentrală

by Spunei Elisabeta

definitivării unei lucrări ce încheie această pregătire. Ideea realizării acestei lucrări nu a apărut din senin, ci a fost o consecinţă a întrebărilor pe care mi le-am pus în timpul licenţei, masteratului, asupra unor fenomene care au loc... more

definitivării unei lucrări ce încheie această pregătire. Ideea realizării acestei lucrări nu a apărut din senin, ci a fost o consecinţă a întrebărilor pe care mi le-am pus în timpul licenţei, masteratului, asupra unor fenomene care au loc în maşinile electrice şi care stau la baza minunatelor transformări energetice din acestea. Am avut şansa să fiu îndrumată către d-l Prof. Univ. Dr. Ing. AUREL CÂMPEANU, membru ASTR, cu care am stabilit tema pe care urma să o dezvolt în teza de doctorat, încă înainte de finalizarea licenţei. Pe toată perioada de pregătire, conducătorul tezei de doctorat s-a implicat cu tact şi cu o grijă părintească în îndeplinirea obiectivului propus. Mi-a fost alături şi mi-a îndrumat paşii pe toată perioada de pregătire, d-l Prof. Univ. Dr. Ing. Ec. ION PIROI. Dânsul, cu devotamentul şi dăruirea proprie, a pus pregătirea mea înaintea altor priorităţi, investind mult timp, energie, urmărind să îmi transfere cât mai mult din cunoştinţele dumnealui. Am cunoscut în timpul elaborării tezei căldura şi devotamentul celor mai apropiaţi formatori. În acest sens sunt recunoscătoare pentru sprijinul de specialitate, moral, acordat de d-l Prof. Univ. Dr. Ing. ION VLAD, care nu şi-a precupeţit nici măcar zilele libere, acceptând să mă îndrume către aplicarea celor mai eficiente metode de optimizare a generatorului sincron. Nu uit, de asemenea, răbdarea cu care d-l Prof. Univ. Dr. Ing. SORIN ENACHE a ascultat prezentările periodice făcute în cadrul etapelor de pregătire, la început mai stângace, corectând anumite inadvertenţe. Şi-a făcut timp pentru a-mi lămuri anumite fenomene d-l Academician TOMA DORDEA, care m-a însufleţit de câte ori am apelat la domnia sa pentru a lămuri anumite fenomene şi a interpreta anumite înregistrări. Am fost îndrumată în investigaţiile practice, şi nu numai, de d-l Prof. Univ. Dr. Ing. GHEORGHE LIUBA, care şi-a sacrificat multe ore pentru a corecta anumite erori care s-ar fi putut strecura în perioada de elaborare a tezei. Tuturor le mulţumesc deodată, dar în mod diferit fiecăruia, pentru tot ce au făcut în formarea mea din această etapă a vieţii. Nu pot uita, de asemenea, sprijinul acordat de specialiştii standului de încercări din UCMR şi a d-lui Director Ing. DAN NICOLAE OBĂDĂU, care mi-a aprobat efectuarea încercărilor, cărora le mulţumesc. Aduc mulţumiri cadrelor didactice de la Facultatea de Inginerie Electrică din Craiova şi de la Facultatea de Inginerie Electrică şi Informatică din Reşiţa, care mi-au fost alături, m-au îndrumat şi au crezut în mine. Mi-au fost alături părinţii şi familia mea, foştii colegi de la CFR şi în special de la fosta Secţie CT3 Caransebeş, îndrumătorii mei din conducerea Diviziei Instalaţii Timişoara şi din Direcţia Instalaţii Bucureşti, cărora le mulţumesc călduros. Pentru a nu mă repeta, mulţumesc membrilor comisiei de evaluare a tezei, care şi-au făcut timp pentru a studia şi a evidenţia anumite aspecte din aceasta. Şi acum, în mod special, mulţumesc d-nei Rector Prof. Univ. Dr. Ing. DOINA FRUNZĂVERDE care mi-a acordat un sprijin moral deosebit, şi nu numai, creându-mi condiţii optime pentru această pregătire. La fiecare întâlnire ocazională sau planificată, domnia sa a avut mereu un zâmbet încurajator pentru mine. Reşiţa, la 07 noiembrie 2012 Cu deosebită consideraţie, Ing. Elisabeta ERINA (SPUNEI)

Lumped-Parameter Network Thermal Analysis of Permanent Magnet Synchronous Motor

2014, Journal of Materials Science Forum, vol. 792, pp. 233-238

This paper develops a model based on an original lumped-parameter network configuration for the thermal transient analysis of a permanent magnet synchronous motor (PMSM). The specific PMSM was designed and optimized for a demanding... more

Fig. 1. Prototype PMSM stator core and winding.

Lumped-Parameter Model for Thermal Transient Analysis

Fig. 6. PMSM temperature distribution, 82.7 s.

Figs. 4 to 7 show the temperature distribution of the PMSM running under maximum load. It was
obtained by a 10-step transient analysis. Analysis time ranges from 0.18 s to 8,233 s and the time-
step ranges from 0.18 s for the first step to 5,626 s for the last step. More specifically, Figs. 4, 5
show the temperature distribution of the stator core and the rotor at the third time-step of the
analysis. Figs. 6, 7 shows the temperature distribution for two steps of the transient analysis. Figs. 6,
7 show that the windings are the critical temperature component of the PMSM.

Fig. 11. Temperature variation of the housing.
Conclusion

Transient heat transfer analysis of housing and PMM using 3–D FE code

2016, Proceedings of the 22nd International Conference on Electrical Machines ICEM 2016, Lausanne, Switzerland, September 4-7

This paper proposes finite element (FE) numerical methodologies for the transient heat transfer analysis of highly integrated actuators for demanding aerospace applications. A FE surface heat source is developed which permits to... more

Unification of anisotropy and FEM-BE models for distribution transformer optimization

2008, Journal of Optoelectronics and Advanced Materials, vol. 10, no. 5, pp. 1143-1148

The paper presents a three-dimensional finite element (3D FEM) anisotropy model, based on a particular scalar potential formulation, for the no load loss evaluation of wound core shell type distribution transformers. The specific 3D FEM... more

Fig. 4. Perspective view of the geometry and the flux density vector plot of a one-phase wound core (B = 1.57 T). Fig. 4 illustrates a perspective view of the geometry, and the flux density vector plot of a wound core, obtained by the 3D FEM anisotropy model. The core is built of high permeability magnetic steel, and the mean flux density used for the FE analysis is 1.57 T. Figs. 5 and 6 display the flux density vector, and nodal plot respectively, of the upper front part of the same wound core, for a mean flux density of 1.66 T.

Fig. 1. Geometry of the FEM-BE model. The use of the hybrid FEM-BE technique results in the accurate evaluation of the magnetic field in the transformer’s active part and in the air region surrounding it, with low computational effort comparing with the FE method. Fig. 1 depicts the 3D one-phase model of the three-phase wound core distribution transformer, consisting of the LV and HV windings and the iron cores. The hybrid FEM-BE model is divided in two subdomains, the transformer’s active part, FEM subdomain, represented by a tetrahedral FE mesh, and the subdomain between the active part and the transformer’s tank walls, BE subdomain, represented by a triangular mesh of its boundaries. 5. Experimental verification of the 3D FEM models

Fig. 3. Detail of the wound core, the excitation coil, and the search coils. Fig. 2. Experimental setup.

Fig. 9. Flux density magnitude distribution along line AB.

Fig. 7. Flux density magnitude distribution along line AB. The computed results, shown in Figs. 7 to 10, verify that the flux density distribution along lines AB and CD is different. Nevertheless, due to the fact that the FE analysis does not take into consideration the stress induced by manufacturing processes, the error of the computed flux density is greater in the inner sheets than in the outer sheets, especially along line AB as can be seen from Figs. 7 and 9.

Fig. 8. Flux density magnitude distribution along line CD.

Fig. 6. Nodal plot of the wound core flux density distribution during no load test (B = 1.66 T). Fig. 5. Vector plot of the wound core flux density distribution during no load test (B = 1.66 T).

Table 2. Three-phase wound core distribution transformer parameters and solution output. 7. Conclusion A 3D FEM anisotropy model for the no load loss evaluation of one-phase and three-phase wound core transformers was developed, based on a computationally efficient scalar potential formulation. The accuracy of the The specific optimization problem presents multiple optima and it was tackled by stochastic optimization algorithms. In [2] the geometric programming was applied, whereas in [5] a heuristic solution was chosen. In the present paper two well known stochastic algorithms were tested the genetic algorithm (GA) and the simulated annealing (SA). Table 2 summarizes the transformer’s parameters and the solution obtained by the SA algorithm, i.e. the most effective of the algorithms tested. The GA was proven to be more computationally expensive as it required a greater number of objective function evaluations, and also the final TOC value was 0.51 % higher than that obtained by the SA algorithm.

Table I. Comparison of the computed and measured no load loss values, of three-phase wound core distribution transformers of various apparent power and working induction ratings. The results obtained by the hybrid FEM-BE technique were validated for a number of three-phase distribution transformers through local leakage field measurements by a Hall Effect probe during the short-circuit test. Also the difference between the measured and the computed short circuit impedance is in most cases less than 2.7 %. Further details on the experimental verification of the hybrid FEM-BE technique may be sought in [3] and [6].

Comparative Analysis of Rotor Losses in High-Speed Permanent Magnet Machines with Different Winding Configurations Considering the Influence of the Inverter PWM

Due to development in fields of materials and power electronics, use of high-speed electrical machines has boosted significantly in recent decades. They are widely used in number of applications, such as electrically assisted... more

FEM package for iron loss evaluation and minimization of two grade lamination wound cores

2008, Journal of Optoelectronics and Advanced Materials, vol. 10, no. 5, pp. 1197-1202

The iron loss and manufacturing cost of a strip-wound core can be reduced by constructing the wound core with combination of average and high magnetization grade steel. The present paper introduces a two-dimensional finite element (2D... more

Design of a Permanent Magnetless Transverse Flux Machine with Rotary Transformer

by Wasi Uddin

This paper presents a new concept of Transverse Flux Machine (TFM) design that uses rotary transformers to replace the Permanent Magnet (PM) field excitation in the rotor. The rotary transformer has an inductive interface that allows for... more

Author's personal copy New model to estimate daily global solar radiation over Nigeria

by Ohiozua Ohijeagbon

and sharing with colleagues.

Design of Initial Topology of Interior Permanent Magnet Synchronous Machine for Hybrid Electric Vehicle

by Mirsad Cosovic

—The design method of initial topology of interior permanent magnet synchronous machine (IPMSM) for hybrid electric vehicle (HEV) propulsion is described in this paper. Design constraints are selected on the basis of limitations imposed... more

Fig. 1. Cross-section of IPMSM with basic geometrical parameters Values of the geometrical parameters that will not be changed in the design process are given in Table I. Dimensions given in Table I are defined on the basis of given constraints and results of analysis conducted for different IPMSM designed to drive HEV [5], [7], [8].

Fig. 2. Flowchart describing selection procedure of the initial design

Fig. 3. Matrix of electrical machine topologies

Results of parametric analysis are presented using three-dimensional diagrams, where each point on_ the diagram represents one machine design. For example, Fig. 4 shows power and current density as a function of design variables computed for different topologies of machine for al —s INRNN aes and ienwepanwm Kattaxeer coanltanes All machine topologies having output power lower than the rated power are eliminated from the initial set of

Fig. 5. Remaining topologies after application of the rated power and maximum current density criteria In an analogous way, analysis for other considered points of torque-speed characteristic is conducted. Then, it is necessary to establish a set of design variations which will contain only those topologies that meet the specified requirements at each considered speed.

Fig. 6. Topologies that satisfy all eliminatory criteria at each speed Machine topologies that satisfy all eliminatory criteria at each speed are shown in Fig. 6.

where is: P, input power from battery pack; Peon power converter losses; Prep, mechanical losses; Prr core losses; Pou copper losses; P2 output power. Fig. 7. Exemple of segmented efficiency map

As an example, efficiency map generated for one geometry, divided into segments, is shown in Fig. 7. — Initial design of electrical machine will be selected based on the values of cost function:

Values of the cost function computed for all design variations are shown in Fig. 9. Design no. 17 (marked with red point on Fig. 9) has the highest value of cost function and it is considered as the best initial design. Fig. 8. Segmented torque-speed plane with HEV’S operating points

Lower and Upper Limits and Increment of Variable Parameters For the selected values of the rotor radius FR. and the stator slot width h,,9, number of turns N,, is changed according to

Design of A Novel Line Start Synchronous Motor Rotor

by Berkan ZÖHRA

2019, Electronics

Line start permanent magnet synchronous motors (LS-PMSM) are preferred more and more in industrial applications, because they can start on their own and because of their high efficiency. In this study, a new LS-PMSM rotor typology is... more

Figure 1. Torque components of the line start permanent magnet synchronous motors (LS-PMSM). Figure 1 shows the torque components of the LS-PMSM. The dotted curve in the figure represents the torque generated by the start cage without a permanent magnet (T;,), and the dashed line represents the braking torque generated by the permanent magnets (Tym). The asynchronous torque (Tasyn zenerated by electromagnetic torque (Te), which is the sum of T; and Tym, is represented by a normal ine. The cage torque (T-) should overcome the braking torque (Tym) generated by the permanent magnets, the load torque (T,), and motor inertia (J;,) in order for the synchronization process in LS-PMSM to be successful.

Figure 2. Optimal LS-PMSM design algorithm. 2.1. Determining the Main Dimensions of the Machine

einai © The slot structures that determine the motor torque characteristic are standardized by NEMA with class A, B, C and D [34]. According to the NEMA standard classes A and B are used in general purpose motors. Also, Class C and D are preferred in applications requiring higher starting torque. As a result a new rotor typology making use of both of the slot types is suggested in this study, to ensure that both the motor start and synchronization performances are high (Figure 3).

Figure 4. Ty = 35 Nm; Jm = 1 p.u. Starting process of the analytical model: (a) speed characteristic and (b) torque characteristic. steady state operation as I, = 9.6 A and n = 91.14%, respectively. When this efficiency value is compared with the 85% value obtained with the standard IM rotor, it can be observed that an efficiency increase of 6.14% has been attained. IE3 efficiency levels have been reached with this efficiency value, obtained using the analytically modelled design without any optimization.

The objective with the optimization studies carried out was to ensure that the motor efficiency exceeds 91.9% (IE4), by determining the proper design parameters. The optimized design parameters can be seen in Figure 5. M; and My in Figure 5 stand for the magnet thickness and width, respectively; Bs0 denotes the slot gap for the rotor upper slots; H; represents the slot depth; and Bs1 denotes the slot diameter. Finally, the rotor diameter (RO) was also included in the optimization studies for determining the optimum air gap value. The parameters used in the optimization study are shown in Table 4 together with their min. and max. values.

The cost function obtained as a result of optimization via the genetic algorithm has been presented in Figure 6 as a graph. As can be seen from the graph, the targeted efficiency (92%) was reached after 383 evolutions. Figure 7 shows the design parameters processed via GA and the distribution of the parameters. It can be understood upon examining the graph that the values of Bs1, Hs, RO, M;, and Mw do not have a homogeneous distribution, and the optimization study has focused more on certain value intervals. In addition, the graph presented indicates that the impact of the Bs0 value on motor performance is quite low. In conclusion, the targeted efficiency was reached as a result of GA optimization at around Bs1 = 8.5 mm, H,; = 1mm, Bs0 = 1.79 mm, M; = 4.48 mm, My = 40.5°, and RO = 61.30 mm. Moreover, the torque ripples generated by the motor have been decreased down to about T,jppje = 7.09 (SI), and the motor rated current was determined as I, = 9.26 A. Table 5 summarizes the change in the design parameters and motor operating characteristic as a result of the GA optimization.

Figure 7. Distribution of genetic algorithm (GA) analysis parameters according to the number of evolutions. The torque ripples in the motor shaft torque resulted in Tripple = 19.98 (SI in the model, whereas this value was decreased to T,ippie = 7.08 (SI) in the new design. Optimization works were carried out for the optimization of this value, which was still quite high in the later stages of the study. The component generated especially in the pole transitions generates ripples in the torque, because of the interactions between the permanent magnets and the stator tooth in LS-PMSM’s [35]. It was observed that when the studies for reducing the ripples in the torque were examined, the focus was on the optimization of the stator tooth and/or permanent magnets, production of the skewed stator, or rotors [36-40].

Figure 8. Ripple and efficiency obtained for different stator skew angles. that the highest efficiency values were obtained as n (skew7) = 91.33%, n(skew8) = 91.29%, and n (skew13) = 91.28%.

Figure 9. Motor synchronization performance for different stator skew angles. As can be seen in Figure 9, showing the impact of s stator skew angle on the motor synchronization performance, the motor has the highest synchronization performance at angles of skew0 and skew8. However, the high torque ripples of around T,jy»1¢ = 7 (SI) are generated for both skew angles, which results in an unacceptable decrease in the motor shaft torque quality. In addition, it was also observed that motor efficiency and performance decreased significantly for skew15 at which a satisfactory decrease in the torque ripple was obtained. In conclusion, skew13 was preferred as the stator skew angle, because of its satisfactory synchronization performance and lowest torque ripples.

Figure 10. The cogging torque generated during a 30° rotation of the motor for skew0 and skew13 skew angles.

Figure 11. Change in electromagnetic torque subject to rotor revolution generated by the motor for skew0 and skew13 angles.

Figure 12. Starting performance for the suggested model with skewed stator under different motor loads. At this stage of the study, the performance characteristic will be examined for the motor design rith a stator having a skew angle of skew13, modelled using the suggested rotor topology. The most nportant problem with LS-PMSM’s is the braking torque generated during operation, because of re permanent magnets. This results in issues such as the LS-PMSM’s becoming too sensitive to ircumstances, such as the overloading of the motor. Figure 12 shows the starting performance under ifferent motor loads for the skewed stator model suggested. As can be seen from the graph, the motor an reach a synchronous speed of 1500 rpm in 0.3 s at T, = 35 Nm (Full Load). An overloading of 14% [, = 40 Nm) results in the failure of the motor synchronization process.

Figure 13. Current and efficiency values obtained from the motor subject to load torque. The graph in Figure 13 shows the current and efficiency values subject to load torque. It can clearly be seen from the graph that the current drawn when the motor is idle is around I, = 2.3 A. This value increases up to I, = 9.84 A at an overloading of 40 Nm. The current drawn at Fi of the load is 2.8 A and 4.3 A at 3 loading. Motor efficiency at + loading increased up to 86% and up to 91% at 3, whereas a maximum value of 91.28% was reached at full load.

Figure 14. The meshed model of the LS-PMSM with suggested rotor design: (a) rotor and (b) stator.

Figure 15. Field reports of the LS-PMSM with suggested rotor design: (a) flux distributions at the stator and rotor cores in steady state operation and (b) vector distribution.

Figure 16. Starting performance under T;, = 35 Nm constant and square load of the suggested model.

Figure 17. Starting performance under different square loads of the suggested model.

Table 1. Main dimensions of the reference motor.

Table 2. Stator design details. The label information for the standard IE2 IMs taken as reference have been given in Table 3.

Table 3. Label information for the reference motor. 2.3. Rotor Design

Table 4. Design parameters used for the optimization of the model.

Table 5. Change in design parameters and motor efficiency after genetic algorithm (GA) optimization. A significant number of analyses were carried out during the optimization studies for reaching an optimum rotor design with a high efficiency. The motor design was evaluated as being two dimensional during the analyses carried out, as it delivers faster solutions, and the analyses were finalized with a Maxwell 2D Transient solver. A Maxwell 3D Transient solver was used during the next stages of the study for a more detailed examination of the model obtained with a high efficiency via GA.

Flux distribution in single phase, Si–Fe, wound transformer cores

2008, ournal of Magnetism and Magnetic Materials, vol. 320, pp. 874-877

This paper shows experimental results of longitudinal flux density and its harmonics at the limb, the yoke and the corner as well as normal flux in the step lap joint of a single phase, Si–Fe, wound transformer core. Results show that... more

Fig. 8. Ratio of normal to longitudinal flux as a function of reference core flux density.

Maximum flux values in the step lap joint of the wound core under test

Fig. 4. Flux density distribution across the corner of the core.

Fig. 2. Flux density distribution across the limb of the core.

Fig. 3. Flux density distribution across the yoke of the core.

Fig. 5. Harmonics components of the localized flux density when the core is magnetized at 1.7, 1.6 and 1.5T.

Fig. 6. Harmonics components of the localized flux density when the core is magnetized at 1.4, 1.3 and 1.0T.

Fig. 7. Detail of a wound core section showing the locations of search coils used for normal flux density measurements (figure not in scale).

Fig. 1. Wound core modeled (dimensions in mm).

3–D FEM and Lumped–Parameter Network Transient Thermal Analysis of Induction and Permanent Magnet Motors for Aerospace Applications

2016, Journal of Materials Science Forum, vol. 856, pp. 245-250

Three dimensional (3–D), finite–element (FE) models and original lumped–parameter networks are developed for the transient thermal analysis of a permanent magnet motor (PMM) and an induction motor (IM) specifically designed and optimized... more

Multiple Grade Lamination Wound Core: A Novel Technique for Transformer Iron Loss Minimization Using Simulated Annealing With Restarts and an Anisotropy Model

2008, IEEE Transactions on Magnetics, vol. 44, no. 6, pp. 1082-1085

This paper introduces a novel technique for iron loss minimization of wound core transformers. The proposed technique involves the evaluation of appropriate design variables of wound cores constructed by a combination of standard and high... more

Design and characteristic analysis of 10 MW class superconducting wind turbine generators with different types of stator and rotor configurations

by Hiroyuki Ohsaki and

2013, H. Ohsaki, L. Quéval, Y. Terao, "Design and characteristic analysis of 10 MW class superconducting wind turbine generators with different types of stator and rotor configurations," 4th International Conference on Clean Electrical Power (ICCEP 2013), Alghero, Italy, June 2013.

Wind power generation facilities have been installed in the world. Recently offshore wind energy attracts greater attention, and development of wind turbine generators of higher power rating becomes more important. Superconducting... more

Fig. 2 Transition of rotor diameter and capacity of wind turbines. Fig. | Global annual and cumulative installed wind capacities.

machines even in partial load conditions. High current density enables high magnetic field and high torque generation, smaller size and lighter weight. Using such characteristics of superconductors we can design higher- output and direct-drive synchronous generators. The important high-temperature superconductors are Bi,Sr,Ca,Cu30;9 (or Bi2223) and YBa,Cu;07. (or YBCO) [8], [9]. High-temperature superconductors such as Bi2223 and YBCO have much higher critical temperature than low-temperature superconductors such as NbTi and Nb;Sn. By the use of superconducting technology we can expect that the mass of nacelle, hub and blades of a 10 MW class superconducting generator can be almost similar to that of 5 MW class conventional machines.

Fig. 5 Fundamental rotor structures with superconducting field coils. Fig. 4 Fundamental stator structures with armature windings.

Fig. 3. A wind power generator system configuration for a direct-drive synchronous machine. Table I indicates the fundamental specifications of 10 MW superconducting wind turbine generators designed in this study. Although there are also some designs using a medium speed gear box even for large capacity wind turbines, we assumed that a direct-drive synchronous generator would be more suitable for a 10 MW class wind turbine. A 10 MW wind turbine has a 160 — 180 m diameter rotor, and a direct-drive synchronous generator rotates at as low as 10 rpm. Therefore, a generator output has a very low frequency. It is transformed into 50 or 60 Hz AC by an AC/DC/AC power converter as shown in Fig. 3.

and are strongly saturated. High electromagnetic force and iron losses should be carefully examined in the design of the machines with slotted iron core. There are three rotor structures. Namely, air-core field coils without iron core shown in Fig. 5 (a), air-core field coils with back iron in Fig. 5 (b), and field coils with salient- pole iron core in Fig. 5 (c). It is an important issue how to use iron core in the stator and rotor sides to reduce the generator size and the superconducting wire length.

Fig. 6 An example of a cross-section structure of a superconducting wind turbine generator. It has reduced the machine size and weight, namely, the stator diameter is 4.7 m and the active mass is 110 tons. However, the NS-SCG needs 1,240 km YBCO superconducting wire, which would cost a lot. Due to the non-salient-pole structure, the air-gap armature windings, and the larger air gap length, the three-phase output voltages are almost sinusoidal as shown in Fig. 8.

Fig. 7 FEM analysis results of flux density distribution and flux lines of the NS-SCG.

Fig. 8 Phase voltage waveforms of three-phase output voltages of the NS-SCG.

FEM ANALYSIS RESULTS OF FOUR-TYPES OF 10 MW WIND TURBINE GENERATORS TABLE II

Optimization of the Electromagnetic Characteristics of a 3-Phase Squirrel-Cage Induction Motor Using FEM

by Konstantinos N. Gyftakis and

The induction motor due to its characteristics forms the most common choice of industrial applications worldwide. In this work, with the use of FEM analysis, a specific induction motor is optimized under three different criteria. The... more

Finite element transient thermal analysis of PMSM for aerospace applications

2012, Proceedings of the 20th International Conference on Electrical Machines ICEM 2012, Marseille, France

This paper describes finite element techniques that reduce the computational cost of the 3D transient thermal analysis. Based on those techniques a finite element package is assembled. The package is used for the transient thermal... more

Fig. 21. Comparison of simulated versus measured winding ends motor temperature variation over 28 min (ambient 25 °C). Fig. 20. Comparison of simulated versus measured housing motor temperature variation over 28 min (ambient 25 °C).

Fig. 22. Comparison of simulated versus measured housing motor temperature variation over 9 min (ambient 70 °C).

Fig. 2. Assembled prototype PMSM on the test rig.

Fig. 3. Cross section of reduced 3D FE model.

Fig. 4. Structure of the developed FE package.

Fig. 6. Thermocouple arrangement for temperature measurements.

Fig. 5. Prototype PMSM inside the thermal chamber.

Fig. 7. 3D FE mesh of the PMSM consisting of 2:10° nodes and 1.2:10 elements. Fig. 8. Cross section of the 3D FE mesh of the PMSM.

Fig. 12. PMSM rotor temperature distribution (2.54 s). Fig. 11. PMSM stator temperature distribution (2.54 s). the best compromise between accuracy and computational cost. The corresponding computational cost for a 10-step transient analysis was 1,523 s and the relative error was 2.3%. For the analyses a typical desktop PC was used (processor: 3.0 GHz Intel Core 2 Duo, memory: 2 GB DDR2).

Fig. 19. Local maximum and local minimum temperature variation over 140 min.

Fig. 10. PMSM stator core temperature distribution (2.54 s). Fig. 9. PMSM winding temperature distribution (2.54 s).

Fig. 16. PMSM temperature distribution (825 s). Fig. 13. PMSM temperature distribution (26.1 s).

Fig. 15. PMSM temperature distribution (261 s). Fig. 14. PMSM temperature distribution (82.7 s).

Fig. 17. PMSM temperature distribution (2607 s).

Fig. 18. PMSM temperature distribution (8233 s).

Design of skewed mounted permanent magnet synchronous generators based on 2D and 3D finite element techniques

2005, Journal of Materials Processing Technology, vol. 161, pp. 288-293

This paper investigates the effects of rotor skew on the performance of permanent magnet synchronous generators. The main effects are concerning reduction of the electromotive force induced, stator current harmonics and torque ripple. The... more

Fig. 1. One pole part of the permanent magnet machine constructed (unskewed rotor): (a) employed triangular mesh; (b) field distribution at no load; (c: magnetic field distribution at full load.

Fig. 2. No load voltage of permanent magnet generator without rotor skew (2D fem simulation).

Fig. 3. No load voltage of permanent magnet generator without rotor skew (experiment).

Fig. 5. Full load voltage of permanent magnet generator without rotor skew (2D fem simulation). Fig. 6. Full load voltage of permanent magnet generator without rotor skew (experiment).

Fig. 4. No load voltage of permanent magnet generator with rotor skew (2D fem simulation).

Fig. 7. Full load voltage of permanent magnet generator with rotor skew (2D fem simulation).

Fig. 10. No load distribution of air-gap radial magnetic flux density (3D fem simulation): (a) without rotor skew; (b) with rotor skew. Fig. 9. Flux density distribution in the air gap in the middle of the rotor (3D and 2D fem simulation).

Analysis of PM Transverse-Flux Outer Rotor Machines with Different Configuration

by Rupert Gouws

2017, IEEE Transactions on Industry Applications

This paper presents electromagnetic analysis of two permanent magnet transverse flux outer rotor machines with and without magnetic shunts. The research started with designing and analysing a permanent magnet transverse flux machine with... more

Application of pseudo-cooling boundary conditions combined with electromagnetic and thermal weak coupling for the analysis of highly integrated aerospace actuators

2017, Proceedings of the 11th IEEE International Symposium on Diagnostics for Electric Machines, Power Electronics and Drives SDEMPED 2017, Tinos, Greece, August 29 - September 1

Two original numerical techniques are presented for the finite–element (FE) unsteady analysis of permanent magnet synchronous motors (PMSM) and induction motors (IM) operating in demanding aerospace applications. The first technique is... more

Dual Rotor Mutually Coupled Switched Reluctance Machine for Wide Speed Operating Range

by Wasi Uddin

A new dual rotor mutually coupled switched reluctance machine (DR-MCSRM) is proposed. The proposed machine has a magnetic circuit similar to the full pitched mutual coupled switched reluctance machine (FP-MCSRM) without short end... more

Investigation of the Impact of the Operational Temperature on the Performance of a Surface Permanent Magnet Motor

2011, Journal of Materials Science Forum, vol. 670, pp. 259-264

This paper investigates the impact of different operational temperatures on the performance of a surface permanent magnet (SPM) machine with neodymium-iron-boron (NdFeB) magnets for electric drive applications. The field distribution of... more

Special air-gap element for 2-D FEM analysis of electrical Machines accounting for rotor skew

2005, IEEE Transactions on Magnetics, vol. 41, no 5, pp. 2020-2023

The paper develops a special air-gap element based on the analytical solution of Laplace equation combined with standard finite elements, for analysis of electrical machines with rotor skew. It has been implemented for the analysis of a... more

Fig. 1. Realistic skewing representation in 3-D model.

Fig. 2. Skewing conception of 2-D multislice model. proportionally with the number of slices. That is why special spectral analysis for interface conditions in the air gap has been proposed, in order to reduce the number of necessary slices to be considered [11].

Fig. 3. One pole part of the permanent magnet machine constructed. a: Field distribution at no load. b: Field distribution at full load.

Fig. 5. No load voltage of permanent magnet machine without rotor skew (measurement). Fig.4. No load voltage of permanent magnet machine without rotor skew (2-L FEM one slice simulation).

Fig.6. Vector potential distribution in the air gap for five slices and mean value computed by the method proposed.

Fig. 7. Flux density distribution in the middle of the air gap at no load (comparison of 2-D and 3-D FEM simulations).

Fig. 8. No load distribution of air-gap radial magnetic flux density (3-D FEM simulation, a: without rotor skew, b: with rotor skew).

Fig. 9. Magnetic field distribution at no load (3-D FEM simulation) a: without rotor skew. b: with rotor skew.

A Computational Efficient Algorithm for the Design of a Line Start Synchronous Motor with Multi-Segment Magnet Rotor

by Yannis L Karnavas and

Line start permanent magnet synchronous motors present many advantages therefore are considered as an attractive choice for being induction motors alternatives. In this context, this paper presents an analytical though simple algorithmic... more

Electromagnetic Finite Element Analysis in Design of Electrical Machines

by GOURAB GHOSH

Finite element analysis or FEA is a very powerful computational technique that has revolutionised design engineering in recent times. Here I intend to talk about its usefulness in design of Electrical Machines. Consider a region of space... more

Figure 1: In a practical Electrical Machine, the structure is so complicated that it is almost impossible to solve the field equations by conventional methods. Finite element analysis or FEA is a very powerful computational technique that has rev: olutionised design engineering in recent times. Here I intend to talk about its usefulness ir design of Electrical Machines. Consider a region of space and a set of dependent variable: which obeys a set of differential equations over that region. Suppose, that region is enclosec by a boundary and the values of those variables at the boundary points are known. Nov consider a mathematical problem that asks to compute the values of those variables at every single point of that region. These kind of problems are called boundary value problems ir mathematics. These are also known as field problems and the set of differential equation: governing the system are called field equations. This is the exact problem that an enginee: faces while attempting to design an electrical machine. The field equations here are nothing but the Maxwell’s Equations and the variables of interest here are magnetic field intensiti and flux density. Once these quantities are known, all other quantities like flux linkage: of different coils, voltages induced, torque and power produced can be easily derived fron simple mathematical relations.

To solve this problem one has to solve the set of differential equations and the expected answer would be a set of mathematical expressions for the set of desired variables in terms of the position co-ordinates over the region under consideration. However a practical electrical machine has a very complicated structure. It is also made out of many different materials like soft and hard magnetic materials, conductor and insulator materials and they have very different set of electromagnetic properties. So the coefficients of the field equations also changes in different regions depending on the properties of the material that is present there. Hence it is nearly impossible to solve those equations by conventional methods. That is where finite element method comes in. Finite element analysis is a method of getting an approximate solution for a boundary value problem. Simply stated, it is a numerical method of solving a set of differential equations at some discrete points over a region in space. Every complicated structure can

Effective Energy Conservation for the Agricultural Sector using Line Start Permanent Magnet Synchronous Motors

by Gaurav Singh

—The growing demand for electricity in developing countries calls for a continuous increase in the installed capacity. However, with installed capacity unable to meet the pace of increase in demand, increase in efficiency of energy... more

Analytical Flux Linkage and EMF Calculation of a Transverse Flux Machine

—Transverse Flux Machines (TFM) exhibit 3-D flux patterns. Their analysis and design are generally based on 3-D Finite Element Methods (FEM). Previous analytical models were mostly limited to Magnetic Equivalent Circuits (MEC). The... more

Fig. 1: A segment with 5 poles of a transverse flux machine (linear representation due to large diameter) M.F.J. Kremers, J.J.H. Paulides, J.L.G. Janssen, E.A. Lomonova

Fig. 2: Geometrical parameters as used in the TFM analysis infinite permeability is valid, since there is no to minimal saturation in the machine without excitation of the coils.

Fig. 3: Illustration of the method of images for magnets fully covered by the stator core The method of images for an aligned position of the magnet array and stator core is shown in Figure 3 for one image in each boundary. These images are virtual regions with magnetization placed at the same distance at the opposite side of the boundary and are indicated as tiled regions in Figure 3. The stator core is replaced by air. In the magnetic charge model three images in each boundary are used. Adding a fourth image in each boundary hardly improves the accuracy as it changes the resulting magnetic flux density with less than 1%.

In this position the stator core and magnets on the mover are in aligned position. Figure 4 (b) shows the difference between the results obtained with the magnetic charge model and FEM. The difference is off course mainly present at the edges of the stator tooth. The maximum difference between the two results is 0.05 T, respectively less than 10%. This difference mainly results due to the assumption of infinite long iron boundaries in the method of images. As a result, flux focussing towards the core is not modeled [17].

Fig. 5: Illustration of the method of images for magnets partially covered by the stator core a mover position of y = —20 mm the magnets in the air gap are partially covered by the stator core. Figure 6 (a) shows the flux density along the z-axis obtained with the charge model. Figure 6 (b) shows a maximum difference with the results obtained by FEM of 0.1 T, respectively less than 20%. For the unaligned position this difference is higher as the flux focussing of the parts of the magnets which are not covered is not modeled accurately. The method of images does not take these parts into account, resulting in an underestimation of the magnetic flux density in the air gap.

Fig. 6: Magnetic flux density along the z-axis for y = —20 mm (a) calculated with the charge model (b) difference between the charge model and FEM

Fig. 7: Flux integration surfaces used to calculate the armature leakage flux

Fig. 8: Position of the cross sections in the core where the flux is calculated

Fig. 10: The setup for measurement of the EMF of the core with a length of 10 mm Fig. 9: Flux components in the z-direction calculated with FEM and Charge Model at several heights in a single core

Fig. 11: EMF of a single core with a width of (a) 10 mm, (b) 26 mm

TABLE I: TABLE WITH PARAMETERS OF THE TFM

DC Excited Flux-Switching Motor: Rotor Structural Optimization

— This paper investigates the effect of various rotor structures on the torque production capability of DC excited flux-switching motors. In these machines two set of windings, excited with DC and three-phase AC currents, are placed... more

Three-Dimensional Analytical Model of Helical Winding PM Machines Including Rotor Eddy-Currents

Helical windings (or zigzag windings) are used in a number of applications, however in electrical machines mainly employed in low-power, high-speed PM brushless DC machines due to the cost-effectiveness of the winding type while... more

Evaluating hybrid optimization algorithms for design of a permanent magnet generator

by Erlend Engevik and

Particle Swarm Optimization (PSO) and Genetic Algorithms (GA) are used to minimize the cost of a permanent magnet (PM) synchronous generator with concentrated windings for tidal power applications. With the use of MATLABs global... more

Design Trends for Line Start Permanent Magnet Synchronous Motors

by Berkan ZÖHRA

2019, 2019 3rd International Symposium on Multidisciplinary Studies and Innovative Technologies (ISMSIT)

Starting from the first years of the proposed period right up to the 2000s, there were commercialization concerns of line start permanent magnet synchronous motors. This situation, which developed as a result of not developing methods to... more

Optimal Placement of Static VAr Compensators (SVC's) Using Particle Swarm Optimization

by farhad honarmand

1-The identification of optimal locations for 3 Static VAr Compensators (SVC's) in an IEEE 30 bus system is considered in this paper. The problem is drafted as an optimization task and the solution is achieved through Particle Swarm... more

Table 1: Parameters of Optimization Techniques In order to validate the proposed optimization strategy to the present problem a dedicated computer program has been developed in MATLAB and extensive simulation results are obtained. The parameters of the optimization algorithm are identified through trial and error methods and these parameters are listed in Table-1.

Figure 1: Circuit diagram of SVC connected to an Infinite Bus. In this study, the SVC is modeled as a variable susceptance that can generate SOMVAr (capacitive mode) or absorb S5OMVAr inductive mode) at nominal (1.0 p.u.) voltage at the desired bus.

The convergence graph of PSO is shown in Figure-2. It can be seen that the graph converges rapidly with number of iterations and attains the minimum value as 0.4408. It is important to mention that the graphs generated are averaged over 15 trial runs. An interesting observation in this work is that about the population size. The population size is generally taken between 10 and 20 in the literature. However such a population did not yield a satisfactory result in this work. This led to a population of 50 particles which guarantees near global optima with least number of iterations. This aspect has been highlighted in Table-2. It also shows the Voltage Stability Index of SVC’s placed at different locations, thus establishing the superiority of PSO with 50 particles.

Peak load sharing based on blade pitch control of wind turbine in the presence of utility supply Peak load sharing based on blade pitch control of wind turbine in the presence of utility supply

by Mansoor Ashraf

The alternate energy resources are replacing conventional energy sources for electricity production because of tremendously increasing environmental pollution and cost of fossil fuels. Wind power is one of the potential alternate energy... more

3-D Numerical Surface Charge Model Including Relative Permeability: The General Theory

One of the still open issues within low-frequency magnetics is the inclusion of µ r in the calculations using the magnetic charge method. In this paper, a new iterative method to consider the relative permeability is investigated. This... more

Fig. 1. | Equivalent magnetic charge distribution on the interface between a material and air.

The variable B, is the remanent flux density. The surface charge density, 0, can now be written as

The self-field, Hi , in the normal direction is calculated by dividing the total surface into infinite smaller sections which all have a constant charge density, 0j,m In the case, where the surface charge is in the x y-plane and the dimensions of the material are (2a x 2b x 2c), U = x —(— la and V = y — (—1)*b. Thus

surface charges, the following boundary conditions are used: The individual surface charge densities are calculated with

The magnetic field created by a surface charge can be calcu- lated by using the following equation:

Rewriting (18) leads to the following equation for the surface charge density on surface i:

As can be seen, o; is dependent on the field generated by the other charges. Since this relationship is nonlinear, it is difficult to calculate the charges directly. Therefore, to calculate the charges, an iterative process is used where o/" is the charge on surface i for the nth iteration. It can be seen that the first iteration results in a constant surface charge density. Therefore, the magnetic field can be calculated analytically for the first iteration. For the second iteration, the surface charge can be calculated analytically; however, this results in a variable charge density, leading, in general, to numerical field calculations. From iteration 3, both the charge and field are calculated numerically.

Fig. 2. Magnetic charge density on the surface of the magnet for iterations 1-5. (a) | iteration. (b) 2 iterations. (c) 4 iterations. (d) 5 iterations. TABLE I

Fig. 3. Magnetic field of a magnet with an artificially high “, of 700 in (a) x-direction and (b) z-direction.

Fig. 4. Magnetic charge distribution of a soft-magnetic block with a relative magnetic permeability of 700 in an external field for the 5th iteration.

Fig. 5. Magnetic field of a soft-magnetic block in an external field with a uy of 700 in (a) x-direction and (b) z-direction.

The Impact of Additive Manufacturing on the Development of Electrical Machines for MEA Applications: A Feasibility Study

by Michele Garibaldi

This paper discusses the potential of Additive Manufacturing (AM) as an innovative means of manufacturing electrical machines, with particular focus on the benefits for the MEA. It is argued that the unrivalled design freedom offered by... more

Fig. 1: Schematic of SLM main components and operating principle. The process starts with the wiper distributing a layer of powder from the supply chamber onto the build platform. A high-power laser beam is then applied selectively on the powder surface, according to the -stl file of the part. The energy fully melts the powders together before the build platform descends by one layer thickness along the z-axis. The process is repeated until completion of the final part.

Special air-gap element for 2D FEM analysis of electrical machines accounting for rotor skew

2004, Proceedings of the 11th Biennial IEEE Conference on Electromagnetic Field Computation CEFC 2004, pp. 357, Seoul, Korea

The paper develops a special air-gap element based on the analytical solution of Laplace equation combined with standard finite elements, for analysis of electrical machines with rotor skew. It has been implemented for the analysis of a... more

Türkiye'de verimli elektrik motorlarına geçiş süreci ve Şebeke Kalkışlı Sabit Mıknatıslı Senkron Motorlar

by Berkan ZÖHRA

2019, International Journal of Multidisciplinary Studies and Innovative Technologies

Elektrik motorları her gelişen ülkede olduğu gibi Türkiye’de de tüketilen elektrikte büyük paya sahiptir. Ayrıca mevcut verilere göre Türkiye elektriğinin %35’lik kısmı sanayide yoğun olarak tercih edilen asenkron motorlar tarafından... more

Elektrik motorları her gelişen ülkede olduğu gibi Türkiye’de de tüketilen elektrikte büyük paya sahiptir. Ayrıca mevcut verilere göre Türkiye elektriğinin %35’lik kısmı sanayide yoğun olarak tercih edilen asenkron motorlar tarafından tüketilmektedir. Bu nedenle bu motorların daha verimli hale getirilmesi ile elektrik tüketiminde de gözle görünür azalma sağlanabilmektedir. Ayrıca büyük oranda asenkron motorların kullanıldığı endüstriyel uygulamalarda daha verimli motorların tercih edilmesi durumunda, firma enerji giderlerinde de tasarruf sağlanabilmektedir. Günümüzde elektrik motorları pazarında çalışma koşullarına cevap verebilecek birçok ürün mevcuttur ve bu ürünler uluslararası standartlar ile karakterize edilmektedir. Elektrik motoru üreticileri tarafından kabul gören bu standartlar ile piyasaya sunulan elektrik motorlarının çerçeve boyutları ve çalışma şartları gibi birçok konuda düzenleme yapılmaktadır. Yoğun olarak kabul gören IEC 60034-30-1 elektrik motoru standartlarına göre 0.12 – 1000 kW mil gücü aralığındaki elektrik motorları verimliliklerine göre; IE1 - Standart Verimlilik, IE2 - Yüksek Verimlilik, IE3 - Premium Verimlilik ve IE4 - Süper Premium Verimlilik şeklinde sınıflandırılmaktadır. Asenkron motorlar teknolojik sınırlamalar ve malzeme limitleri neticesinde IE3 ve altı verimlikte piyasa sunulabilmektedir. Buna karşın asenkron motor rotorunun yüksek performanslı sabit mıknatıslar kullanılarak modellenmesi ile yüksek verim, güç faktörü ve güç yoğunluğu sunabilen elektrik motorları üretilebilmektedir. Rotorunda sincap kafesi ve sabit mıknatısların bir arada kullanılması neticesinde hibrit bir yapıya sahip olan bu motorlar literatürde “Şebeke Kalkışlı Sabit Mıknatıslı Senkron Motor” olarak yer almaktadır. Asenkron motorlar gibi sağlam ve az bakım gerektiren bu motorlar, rotorunda yer alan sabit mıknatıslar sayesinde IE3 üstü standartlarda verimlilik sunabilmektedir. Diğer bir yandan bu motorların üretim süreci sabit mıknatıs nedeni ile asenkron motorlara göre ile daha karmaşık ve maliyetli olmaktadır. Yapılan bu çalışma ile üretim maliyeti yüksek olan şebeke kalkışlı sabit mıknatıslı senkron motorların Türkiye pazarına giriş ücretleri ve yüksek verimli çalışma koşullarında motorun ne kadar zamanda alım maliyetini amorti edeceği tartışılmıştır. Ayrıca Türkiye’deki verimli motorlara geçiş sürecine ışık tutulmuştur.