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Fig. 13. Top view of the MIM transmission line. Fig. 12. Measured parameters of the transmission line. (a) Attenuation constant. (b) Phase constant. (c) Characteristic impedance. (d) Quality factor.

Figure 13 Top view of the MIM transmission line. Fig. 12. Measured parameters of the transmission line. (a) Attenuation constant. (b) Phase constant. (c) Characteristic impedance. (d) Quality factor.

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generally not so accurate across a wide frequency range. How- ever, a baseband phase rotator is very helpful to compensate the I/Q imbalance.
generally not so accurate across a wide frequency range. How- ever, a baseband phase rotator is very helpful to compensate the I/Q imbalance.
TARGET SPECIFICATION
TARGET SPECIFICATION
Fig. 2. Increment of a required CNR by RF impairments (ACNR) to achieve a PER of 10-3.
Fig. 2. Increment of a required CNR by RF impairments (ACNR) to achieve a PER of 10-3.
Fig. 4. Block diagram of the 60-GHz quadrature local synthesizer. Fig. 3. Block diagram of the 60-GHz direct-conversion transceiver.
Fig. 4. Block diagram of the 60-GHz quadrature local synthesizer. Fig. 3. Block diagram of the 60-GHz direct-conversion transceiver.
Fig. 5. Schematic of the quadrature injection-locked oscillator. (a) Quadrature injection-locked oscillator, (b) Capacitor part.
Fig. 5. Schematic of the quadrature injection-locked oscillator. (a) Quadrature injection-locked oscillator, (b) Capacitor part.
Fig. 6. Simulated locking range.
Fig. 6. Simulated locking range.
The LNA output is separated into I and Q paths through a par- allel-line transformer and each is connected to a down-conver- sion mixer. A 1-dB 7-type attenuator is placed after the LNA to reduce the reflected signal. Fig. 10 shows a circuit schematic of the down-conversion mixer, which is a kind of passive mixer. In this design, a parallel-line transformer is used for single-to-dif- ferential conversion. Since a transformer balun generally causes an imbalance in differential signals, a differential amplifier that has a high common-mode rejection ratio (CMRR) is used to re- duce the imbalance as explained in the following equations: where & and 6 are gain and phase mismatches in input differen- tial signal, and a is CMRR. Output differential signal (Aout+, Aout—) has less mismatch than input differential signal (Ajn4, Ajn—) due to the common-mode rejection. For example, when CMRR is 10 dB, gain and phase mismatches in a balun can be
The LNA output is separated into I and Q paths through a par- allel-line transformer and each is connected to a down-conver- sion mixer. A 1-dB 7-type attenuator is placed after the LNA to reduce the reflected signal. Fig. 10 shows a circuit schematic of the down-conversion mixer, which is a kind of passive mixer. In this design, a parallel-line transformer is used for single-to-dif- ferential conversion. Since a transformer balun generally causes an imbalance in differential signals, a differential amplifier that has a high common-mode rejection ratio (CMRR) is used to re- duce the imbalance as explained in the following equations: where & and 6 are gain and phase mismatches in input differen- tial signal, and a is CMRR. Output differential signal (Aout+, Aout—) has less mismatch than input differential signal (Ajn4, Ajn—) due to the common-mode rejection. For example, when CMRR is 10 dB, gain and phase mismatches in a balun can be
Fig. 10. Schematic of the down-conversion mixer. sistor. Thus, the common-mode rejection (CMR) is realized in the matching blocks of differential amplifiers. The shunt part in the matching block works as a short stub for a differential-mode signal because it forms a virtual ground. For a common-mode signal, it works as an open stub. However, the stub parasitics reduce the impedance at the virtual ground node even for a improved from 1.0 dB and 5° to 0.10 dB and 0.49°, respectively. Thus, a high CMRR amplifier is required before a double-bal- anced mixer to improve the dynamic range of receiver, e.g., IM2. - —_= eS ee The common differential amplifier using a tail transistor can not maintain a high CMRR at millimeter-wave frequency be- cause of the parasitic capacitance at the drain of the tail tran-
Fig. 10. Schematic of the down-conversion mixer. sistor. Thus, the common-mode rejection (CMR) is realized in the matching blocks of differential amplifiers. The shunt part in the matching block works as a short stub for a differential-mode signal because it forms a virtual ground. For a common-mode signal, it works as an open stub. However, the stub parasitics reduce the impedance at the virtual ground node even for a improved from 1.0 dB and 5° to 0.10 dB and 0.49°, respectively. Thus, a high CMRR amplifier is required before a double-bal- anced mixer to improve the dynamic range of receiver, e.g., IM2. - —_= eS ee The common differential amplifier using a tail transistor can not maintain a high CMRR at millimeter-wave frequency be- cause of the parasitic capacitance at the drain of the tail tran-
Fig. 11. Structure of the transmission line. (a) Simplified TL structure. (b) Manual dummy (proposed design). (c) Auto dummy (common design).
Fig. 11. Structure of the transmission line. (a) Simplified TL structure. (b) Manual dummy (proposed design). (c) Auto dummy (common design).
Fig. 14. Measured and modeled characteristic impedance of the MIM trans- mission line.
Fig. 14. Measured and modeled characteristic impedance of the MIM trans- mission line.
POWER CONSUMPTION SUMMARY implemented on a 4.2-mm x 4.2-mm chip. The core areas of the transmitter and the receiver including all matching blocks are 3.5 mm? and 3.8 mm?, respectively. The 20-GHz PLL is implemented on a separated chip [15], [20], and the core area of the PLL is 1.2 mm?. The same 20-GHz PLL as in [15] is used. In the measurement, two test boards are used as Tx and Rx, and the 20-GHz PLL on a probe station provides the injection signal for the boards. The transmitter, receiver, and PLL consume 186 mW, 106 mW, and 66 mW, respectively, as summarized in Table I. The transmitter and PLL use a 1.2-V supply, and the receiver uses a 1.0-V supply.
POWER CONSUMPTION SUMMARY implemented on a 4.2-mm x 4.2-mm chip. The core areas of the transmitter and the receiver including all matching blocks are 3.5 mm? and 3.8 mm?, respectively. The 20-GHz PLL is implemented on a separated chip [15], [20], and the core area of the PLL is 1.2 mm?. The same 20-GHz PLL as in [15] is used. In the measurement, two test boards are used as Tx and Rx, and the 20-GHz PLL on a probe station provides the injection signal for the boards. The transmitter, receiver, and PLL consume 186 mW, 106 mW, and 66 mW, respectively, as summarized in Table I. The transmitter and PLL use a 1.2-V supply, and the receiver uses a 1.0-V supply.
Fig. 16. Measured characteristics of the parallel-line transformer. (a) Ampli- tude and phase differences. (b) Insertion loss. Fig. 15. Structure of the parallel-line transformer.
Fig. 16. Measured characteristics of the parallel-line transformer. (a) Ampli- tude and phase differences. (b) Insertion loss. Fig. 15. Structure of the parallel-line transformer.
Fig. 17. Die micrograph of the transceiver and receiver.
Fig. 17. Die micrograph of the transceiver and receiver.
PERFORMANCE SUMMARY OF THE LOCAL SYNTHESIZER TABLE III
PERFORMANCE SUMMARY OF THE LOCAL SYNTHESIZER TABLE III
Fig. 19. Die micrograph of the 60-GHz QILO.
Fig. 19. Die micrograph of the 60-GHz QILO.
! Only power consumption of VCO is used. ? Only power consumption of QILO is used. Fig. 20. Phase noise of 60-GHz QILO and 20-GHz PLL.
! Only power consumption of VCO is used. ? Only power consumption of QILO is used. Fig. 20. Phase noise of 60-GHz QILO and 20-GHz PLL.
Fig. 21. Frequency spectrum for free-running and locked states.
Fig. 21. Frequency spectrum for free-running and locked states.
Fig. 23. Required injection power with process and voltage variations. Fig. 22. Measured locking range.
Fig. 23. Required injection power with process and voltage variations. Fig. 22. Measured locking range.
Fig. 24. Measured conversion gain of the transmitter.
Fig. 24. Measured conversion gain of the transmitter.
TABLE V PERFORMANCE SUMMARY OF THE TRANSMITTER
TABLE V PERFORMANCE SUMMARY OF THE TRANSMITTER
2 PERFORMANCE SUMMARY OF THE RECEIVER Fig. 26. Measured conversion gain of the receiver.
2 PERFORMANCE SUMMARY OF THE RECEIVER Fig. 26. Measured conversion gain of the receiver.
Fig. 27. Measured noise figure of the entire receiver chain. >ERFORMANCE COMPARISON OF 60-GHZ LOCAL SYNTHESIZERS
Fig. 27. Measured noise figure of the entire receiver chain. >ERFORMANCE COMPARISON OF 60-GHZ LOCAL SYNTHESIZERS
driven by a quadrature oscillator, which allows high-quality modulation. nected through RF cables. 20-dB external amplifiers are used to compensate —15 dB cable loss, and the input power of 20-GHz incident signal to QILO is about +3 dBm in this measurement. The maximum communication distance is about 2.7 m for BPSK and QPSK within a BER of <1073, and it is about 0.2 m for 8PSK and 16QAM. The minimum communication distance is about 5 mm determined by the limitation of PCB. The low-gain mode of LNA is used for short-distance receiving. When a wider bandwidth is used, the data rate can be extended up to 8 Gb/s in QPSK mode and 11 Gb/s in 16QAM mode within a BER of <1073 and a 25% roll-off. Table VIII summarizes the measure- ment results at the maximum data rates, which are limited by the baseband bandwidth and degradation of RF gain flatness.
driven by a quadrature oscillator, which allows high-quality modulation. nected through RF cables. 20-dB external amplifiers are used to compensate —15 dB cable loss, and the input power of 20-GHz incident signal to QILO is about +3 dBm in this measurement. The maximum communication distance is about 2.7 m for BPSK and QPSK within a BER of <1073, and it is about 0.2 m for 8PSK and 16QAM. The minimum communication distance is about 5 mm determined by the limitation of PCB. The low-gain mode of LNA is used for short-distance receiving. When a wider bandwidth is used, the data rate can be extended up to 8 Gb/s in QPSK mode and 11 Gb/s in 16QAM mode within a BER of <1073 and a 25% roll-off. Table VIII summarizes the measure- ment results at the maximum data rates, which are limited by the baseband bandwidth and degradation of RF gain flatness.
Fig. 28. Measured modulated spectra at Tx output. (a) BPSK, (b) QPSK, (c) 8PSK, (d) 16QAM.
Fig. 28. Measured modulated spectra at Tx output. (a) BPSK, (b) QPSK, (c) 8PSK, (d) 16QAM.
for 8PSK and 16QAM, which was evaluated with in-package antennas called the post-wall waveguide aperture antenna (PWAA). The maximum data rates are 8 Gb/s in QPSK mode and 11 Gb/s in 16QAM mode within a BER of < 1073. Tx, Rx, and PLL consume 186 mW, 106 mW, and 66 mW, respectively.
for 8PSK and 16QAM, which was evaluated with in-package antennas called the post-wall waveguide aperture antenna (PWAA). The maximum data rates are 8 Gb/s in QPSK mode and 11 Gb/s in 16QAM mode within a BER of < 1073. Tx, Rx, and PLL consume 186 mW, 106 mW, and 66 mW, respectively.
MEASURED CONSTELLATION, DATA RATE, EVM, AND COMMUNICATION DISTANCE AT MAXIMUM DATA RATE PERFORMANCE COMPARISON WITH 60-GHZ TRANSCEIVERS
MEASURED CONSTELLATION, DATA RATE, EVM, AND COMMUNICATION DISTANCE AT MAXIMUM DATA RATE PERFORMANCE COMPARISON WITH 60-GHZ TRANSCEIVERS
Rui Murakami received the B.E. and M.E. degrees in physical electronics from Tokyo Institute of Tech- nology, Tokyo, Japan, in 2009 and 2011, respectively. He is currently with Panasonic Corporation, Japan.
Rui Murakami received the B.E. and M.E. degrees in physical electronics from Tokyo Institute of Tech- nology, Tokyo, Japan, in 2009 and 2011, respectively. He is currently with Panasonic Corporation, Japan.
Win Chaivipas (S’06-M’10) received the B.S. degree in electrical engineering from Sirindhorn International Institute of Technology Pathumthani Thailand, the M.S. degree from Royal Institute of Technology Stockholm, Sweden, and the Ph.D. degree from Tokyo Institute of Technology, Japan, in 2001, 2003, and 2009, respectively. Tis eee pee ee es A eT
Win Chaivipas (S’06-M’10) received the B.S. degree in electrical engineering from Sirindhorn International Institute of Technology Pathumthani Thailand, the M.S. degree from Royal Institute of Technology Stockholm, Sweden, and the Ph.D. degree from Tokyo Institute of Technology, Japan, in 2001, 2003, and 2009, respectively. Tis eee pee ee es A eT
Related topics:
Next Generation NetworksBit Error RateSolid State Devices and CircuitsFrequency synthesizersPhase NoiseElectrical and Electronic EngineeringPhase Locked LoopsQuadrature Amplitude ModulationPhase Lock LoopInjection Locking
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